LTC4012IUF-3#PBF - LINEAR TECHNOLOGY

40123fb

LTC4012-3

Typical applicaTion

FeaTures

applicaTions

DescripTion

High Efficiency,

Multi-Chemistry Battery Charger

with PowerPath Control

The LTC

4012-3 is a constant-current/constant-voltage

battery charger controller. It uses a synchronous quasi-

constant frequency PWM control architecture that will not

generate audible noise with ceramic bulk capacitors. Charge

current is set by external resistors and can be monitored

as an output voltage across the programming resistor.

With no built-in termination, the LTC4012-3 charges a

wide range of batteries under external control.

The LTC4012-3 features fully adjustable output voltage. For

charge management and safety, the IC includes an input

P-channel MOSFET ideal diode controller, battery (output)

overvoltage protection, reverse charge current protection,

PWM soft-start and robust non-overlap control for an all

N-channel MOSFET PWM power stage.

The device includes AC adapter input current limiting, which

maximizes the charge rate for a fixed input power level. An

external sense resistor programs the input current limit,

and the ICL status pin indicates reduced charge current as

a result of AC adapter current limiting. Ideal diode control

at the adaptor input improves charger efficiency.

The CHRG status pin is active during all charging modes,

including special indication for low charge current.

n General Purpose Battery Charger Controller

n Efficient 550kHz Synchronous Buck PWM Topology

n ±0.5% Output Float Voltage Accuracy

n Programmable Charge Current: 4% Accuracy

n Programmable AC Adapter Current Limit:

3% Accuracy

n No Audible Noise with Ceramic Capacitors

n INFET Low Loss Ideal Diode PowerPath™ Control

n Wide Input Voltage Range: 6V to 28V

n Wide Output Voltage Range: 2V to 28V

n Indicator Outputs for Charging, C/10 Current

Detection and Input Current Limiting

n Analog Charge Current Monitor

n Micropower Shutdown

n 20-Pin 4mm × 4mm × 0.75mm QFN Package

n Notebook Computers

n Portable Instruments

n Battery Backup Systems

Efficiency at DCIN = 20V

CLP

FROM

ADAPTER

13V TO 20V 0.1µF 5.1k

25mΩ

3.01k

0.1µF

6.8µH

20µF

33mΩ

INFET

0.1µF

2µF

0.1µF

6.04k

26.7k

LTC4012-3

DCIN

CHRG

ICL

SHDN

ITH

PROG

CLN

BOOST

TGATE

INTVDD

BGATE

TO/FROM

MCU

GND

CSP

CSN

BAT

FBDIV

VFB

20µF

POWER TO

SYSTEM

4.7nF

32.8k

4012-3 TA01

12.3V

Li-Ion

BATTERY

3.01k

301k +

CHARGE CURRENT (A)

EFFICIENCY (%)

POWER LOSS (mW)

100

1000

10000

0.5 1 1.5 2

4012-3 TA02

2.5 3

LTC4012

LTC4012-3

GND

ACP

PIN 5 NAME

VOUT = 12.3V

RSENSE = 33mΩ

RIN = 3.01k

RPROG = 26.7k

EFFICIENCY

POWER LOSS

PART

L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and

PowerPath and ThinSOT are trademarks of Linear Technology Corporation. All other trademarks

are the property of their respective owners. Protected by U.S. Patents including 5723970.

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LTC4012-3

pin conFiguraTion

absoluTe MaxiMuM raTings

DCIN ............................................................–14V to 30V

DCIN to CLP ................................................–32V to 20V

CLP, CLN or SW to GND ............................. –0.3V to 30V

CLP to CLN ............................................................±0.3V

CSP, CSN or BAT to GND ........................... –0.3V to 28V

CSP to CSN ............................................................±0.3V

BOOST to GND ........................................... –0.3V to 36V

BOOST to SW...............................................–0.3V to 7V

SHDN or VFB to GND .................................... –0.3V to 7V

CHRG or ICL to GND .................................. –0.3V to 30V

Operating Temperature Range

(Note 2) ............................................. –40°C to 125°C

Junction Temperature (Note 3) ............................. 125°C

Storage Temperature Range ..................–65°C to 150°C

(Note 1)

20 19 18 17 16

678

TOP VIEW

UF PACKAGE

20-LEAD (4mm s 4mm) PLASTIC QFN

9 10

CLN

CLP

INFET

DCIN

GND

CSP

CSN

PROG

ITH

BAT

BOOST

TGATE

INTVDD

BGATE

SHDN

CHRG

ICL

VFB

FBDIV

TJMAX = 125°C, JA = 37°C/W

EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB

orDer inForMaTion

LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE

LTC4012CUF-3#PBF LTC4012CUF-3#TRPBF 40123 20-Lead (4mm × 4mm) Plastic QFN 0°C to 85°C

LTC4012IUF-3#PBF LTC4012IUF-3#TRPBF 40123 20-Lead (4mm × 4mm) Plastic QFN –40°C to 125°C

Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.

Consult LTC Marketing for information on non-standard lead based finish parts.

For more information on lead free part marking, go to: http://www.linear.com/leadfree/

For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/

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LTC4012-3

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Charge Voltage Regulation

VTOL VBAT Accuracy (See Test Circuits)

C-Grade

I-Grade

–0.5

–0.8

–1.0

0.5

0.8

1.0

IVFB VFB Input Bias Current VFB = 1.2V ±20 nA

RON FBDIV On Resistance ILOAD = 100µA l85 190 Ω

ILEAK-FBDIV FBDIV Output Leakage Current SHDN = 0V, FBDIV = 0V l–1 0 1 µA

VBOV VFB Overvoltage Threshold l1.235 1.281 1.32 V

Charge Current Regulation

ITOL Charge Current Accuracy with RIN = 3.01k,

6V < BAT < 18V

RPROG = 26.7k

C-Grade

I-Grade

–4

–5

–9.5

9.5

VSENSE = 0mV, PROG = 1.2V –12.75 –11.67 –10.95 µA

AICurrent Sense Amplifier Gain (PROG ∆I) with

RIN = 3.01k, 6V < BAT < 18V

VSENSE Step from 0mV to 5mV,

PROG = 1.2V

–1.78 –1.66 –1.54 µA

VCS-MAX Maximum Peak Current Sense Threshold Voltage

per Cycle (RIN = 3.01k)

ITH = 2V, C-Grade

ITH = 2V, I-Grade

ITH = 5V

140

125

195

325

250

265

430

VC10 C/10 Indicator Threshold Voltage PROG Falling 340 400 460 mV

VREV Reverse Current Threshold Voltage PROG Falling 180 253 295 mV

Input Current Regulation

VCL Current Limit Threshold CLP – CLN

C-Grade

I-Grade

100

103

104

108

ICLN CLN Input Bias Current CLN = CLP ±100 nA

VICL ICL Indicator Threshold (CLP – CLN) – VCL –8 –5 –2 mV

CLP Supply

OVR Operating Voltage Range 6 28 V

VUVLO CLP Undervoltage Lockout Threshold CLP Increasing l4.65 4.85 5.25 V

VUV(HYST) UVLO Threshold Hysteresis 200 mV

ICLPO CLP Operating Current CLP = 20V, No Gate Loads 2 3 mA

Shutdown

VIL SHDN Input Voltage Low l300 mV

VIH SHDN Input Voltage High l1.4 V

RIN SHDN Pull-Down Resistance 40 kΩ

ICLPS CLP Shutdown Current CLP = 12V, DCIN = 0V

SHDN = 0V

l15

350

500

µA

ILEAK-BAT BAT Leakage Current SHDN = 0V or DCIN = 0V,

0V ≤ CSP = CSN = BAT ≤ 18V

l–1.5 0 1.5 µA

ILEAK-CSN CSN Leakage Current SHDN = 0V or DCIN = 0V,

0V ≤ CSP = CSN = BAT ≤ 20V

l–1.5 0 1.5 µA

ILEAK-CSP CSP Leakage Current SHDN = 0V or DCIN = 0V,

0V ≤ CSP = CSN = BAT ≤ 20V

l–1.5 0 1.5 µA

ILEAK-SW SW Leakage Current SHDN = 0V or DCIN = 0V,

0V ≤ SW ≤ 20V

l–1 0 2 µA

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. DCIN = 20V, BAT = 12V, GND = 0V unless otherwise noted. (Note 2)

40123fb

LTC4012-3

elecTrical characTerisTics

The l denotes the specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. DCIN = 20V, BAT = 12V, GND = 0V unless otherwise noted. (Note 2)

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

INTVDD Regulator

INTVDD Output Voltage No Load l4.85 5 5.15 V

∆VDD Load Regulation IDD = 20mA –0.4 –1 %

IDD Short-Circuit Current (Note 5) INTVDD = 0V 50 85 130 mA

Switching Regulator

VACP AC Present Charge Enable Threshold Voltage DCIN – BAT, DCIN Rising

C-Grade

I-Grade

350

300

500

650

700

IITH ITH Current ITH = 1.4V –40/+90 µA

fTYP Typical Switching Frequency 467 550 633 kHz

fMIN Minimum Switching Frequency CLOAD = 3.3nF 20 25 kHz

DCMAX Maximum Duty Cycle CLOAD = 3.3nF 98 99 %

tR-TG TGATE Rise Time CLOAD = 3.3nF, 10% – 90% 60 110 ns

tF-TG TGATE Fall Time CLOAD = 3.3nF, 90% – 10% 50 110 ns

tR-BG BGATE Rise Time CLOAD = 3.3nF, 10% – 90% 60 110 ns

tF-BG BGATE Fall Time CLOAD = 3.3nF, 90% – 10% 60 110 ns

tNO TGATE, BGATE Non-Overlap Time CLOAD = 3.3nF, 10% – 10% 110 ns

PowerPath Control

IDCIN DCIN Input Current 0V ≤ DCIN ≤ CLP l–10 60 µA

VFTO Forward Turn-On Voltage (DCIN Detection Threshold) DCIN-CLP, DCIN Rising l15 60 mV

VFR Forward Regulation Voltage DCIN-CLP l15 25 35 mV

VRTO Reverse Turn-Off Voltage DCIN-CLP, DCIN Falling l–45 –25 –15 mV

VOL(INFET) INFET Output Low Voltage, Relative to CLP DCIN-CLP = 0.1V, IINFET =1µA –6.5 –5 V

VOH(INFET) INFET Output High Voltage, Relative to CLP DCIN-CLP = –0.1V, IINFET =–5µA –250 250 mV

tIF(ON) INFET Turn-On Time To CLP-INFET > 3V, CINFET = 1nF 85 180 µs

tIF(OFF) INFET Turn-Off Time To CLP-INFET < 1.5V, CINFET = 1nF 2.5 6 µs

Indicator Outputs

VOL Output Voltage Low ILOAD = 100µA, PROG = 1.2V 500 mV

ILEAK Output Leakage SHDN = 0V, DCIN = 0V, VOUT = 20V l–10 10 µA

IC10 CHRG C/10 Current Sink CHRG = 2.5V l15 25 38 µA

Note 1: Stresses beyond those listed under Absolute Maximum Ratings

may cause permanent damage to the device. Exposure to any Absolute

Maximum Rating condition for extended periods may affect device

reliability and lifetime.

Note 2: The LTC4012C-3 is guaranteed to meet performance specifications

over the 0°C to 85°C operating temperature range. The LTC4012I-3 is

guaranteed to meet performance specifications over the –40°C to 125°C

operating temperature range.

Note 3: Operating junction temperature TJ (in °C) is calculated from

the ambient temperature TA and the total continuous package power

dissipation PD (in watts) by the formula TJ = TA + (θJA • PD). Refer to the

Applications Information section for details.

Note 4: All currents into device pins are positive; all currents out of device

pins are negative. All voltages are referenced to GND, unless otherwise

specified.

Note 5: Output current may be limited by internal power dissipation. Refer

to the Applications Information section for details.

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LTC4012-3

Battery Load Dump

BATTERY VOLTAGE

(500mV/DIV)

LOAD

STATE

TIME (1ms/DIV)

CLP = 20V

VOUT = 12.3V

4012-3 G06

DISCONNECT

RECONNECT

12.1V 1A

Typical perForMance characTerisTics

Efficiency at DCIN = 20V, BAT = 8V

VFB Line Regulation

CHARGE CURRENT (A)

EFFICIENCY (%)

POWER LOSS(mW)

100

1000

10000

0.5 1 1.5 2

4012-3 G01

2.5 3

POWER LOSS

EFFICIENCY

RSENSE = 33mΩ

RIN = 3.01k

Efficiency at DCIN = 20V, BAT = 12V

CHARGE CURRENT (A)

EFFICIENCY (%)

POWER LOSS(mW)

100

1000

10000

0.5 1 1.5 2

4012-3 G02

2.5 3

95 EFFICIENCY

POWER LOSS

RSENSE = 33mΩ

RIN = 3.01k

Efficiency at DCIN = 20V, BAT = 16V

CHARGE CURRENT (A)

EFFICIENCY (%)

POWER LOSS(mW)

100

1000

10000

0.5 1 1.5 2

4012-3 G03

2.5 3

EFFICIENCY

POWER LOSS

RSENSE = 33mΩ

RIN = 3.01k

CLP (V)

VFB ERROR (%)

0.02

0.06

0.10

4012-3 G04

–0.02

–0.06

0.04

0.08

–0.04

–0.08

–0.10 10 15 20 30

LTC4012-3 TEST CIRCUIT

BAT (V)

RON (Ω)

100

150

175

200

10 20 25

300

4012-3 G05

125

5 15

225

250

275

CLP = BAT + 3V

(CLP ≥ 6V)

FBDIV Pin RON vs Battery Voltage

(TA = 25°C unless otherwise noted. L = IHLP-2525 6.8µH)

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LTC4012-3

Gate Drive Non-Overlap

Charge Current Line Regulation

Input Current Limit

DCIN (V)

–0.5

CHARGE CURRENT ERROR (%)

–0.4

–0.2

–0.1

0.5

0.2

10 15

4012-3 G08

–0.3

0.3

0.4

0.1

20 25

ICHG = 1A

ICHG = 2A

ICHG = 3A

BAT = 6V

RSENSE = 33mΩ

RIN = 3.01k

Charge Current Load Regulation

BAT (V)

11.0

CHARGE CURRENT (A)

3.5

3.0

2.5

2.0

1.5

1.0

0.5

–0.5

12.6

4012-3 G09

11.4 11.8 12.2 13.0

ICHG = 3A

ICHG = 2A

ICHG = 1A

DCIN = 20V

RSENSE = 33mΩ

RIN = 3.01k

SYSTEM LOAD (A)

CURRENT (A)

0.5

1.0

1.5

1.5 2.5

4012-3 G10

–0.5

–1.0 0.5 1.0 2.0

2.0

2.5

3.0

IIN

ICHG

ICL STATE

2.5A BULK CHARGE

2.1A INPUT CURRENT LIMIT

PWM Soft-Start

ICHG

2A/DIV

TIME (500µs/DIV) 4012-3 G11

ITH

1V/DIV

PROG

1V/DIV

SHDN

5V/DIV

TIME (80ns/DIV)

EXTERNAL FET DRIVE (1V/DIV)

4012-3 G12

TGATE

BGATE

Typical perForMance characTerisTics

(TA = 25°C unless otherwise noted. L = IHLP-2525 6.8µH)

Charge Current Accuracy

BAT (V)

–6

CHARGE CURRENT ERROR (%)

–4

–3

–2

108 20 2422

4012-3 G07

–5

642 1412 16 18

–1

DCIN = 12V

RPROG = 26.7k

DCIN = 24V

RPROG = 35.7k

RSENSE = 33mΩ

RIN = 3.01k

40123fb

LTC4012-3

Typical perForMance characTerisTics

PWM Frequency vs Duty Cycle

Battery Shutdown Current

DUTY CYCLE (%)

PWM FREQUENCY (kHz)

100

200

300

400

500

600

20 40 60 80

4012-3 G13

100

CLP = 6V

CLP = 12V

CLP = 20V

CLP = 25V

ICHG = 750mA

PWM Frequency

vs Charge Current

CHARGE CURRENT (A)

PWM FREQUENCY (kHz)

100

200

300

400

600

0.5 1.0 1.5 2.0

4012-3 G14

2.5 3.0

500

BAT = 14.5V

BAT = 12V

BAT = 5V

CLP = 15V

RSENSE = 33mΩ

RIN = 3.01k

BATTERY VOLTAGE (V)

BATTERY CURRENT (µA)

5 10 15

4012-3 G15

20 25

LTC4012-3,

BAT PINS

DCIN = 20V

LTC4012-3,

ALL PINS

DCIN = 0V

DC1256-CLASS

APPLICATION

DCIN = 0V

(TA = 25°C unless otherwise noted. L = IHLP-2525 6.8µH)

INFET Response Time to

DCIN Short to Ground

VGS =0V

TIME (1µs/DIV)

DCIN = 15V

INFET = Si7423DN

IOUT = <50mA

VOUT = 12.3V

COUT = 0.27F

4012-3 G16

PFET VGS (1V/DIV)

IDCIN, REVERSE

(5A/DIV)

40123fb

LTC4012-3

pin FuncTions

CLN (Pin 1): Adapter Input Current Limit Negative Input.

The LTC4012-3 senses voltage on this pin to determine if

less charge current should be sourced to limit total input

current. The threshold is set 100mV below the CLP pin. An

external filter should be used to remove switching noise.

This input should be tied to CLP if not used. Operating

voltage range is (CLP – 110mV) to CLP.

CLP (Pin 2): Adapter Input Current Limit Positive Input.

The LTC4012-3 also draws power from this pin, including

a small amount for some shutdown functions. Operating

voltage range is GND to 28V.

INFET (Pin 3): PowerPath Control Output. This output

drives the gate of a PMOS pass transistor connected

between the DC input (DCIN) and the raw system supply

rail (CLP) to maintain a forward voltage of 25mV when a

DC input source is present. INFET is internally clamped

about 6V below CLP. Maximum operating voltage is CLP,

which is used to turn off the input PMOS transistor when

the DC input is removed.

DCIN (Pin 4): DC Sense Input. One of two voltage sense

inputs to the internal PowerPath controller (the other input

to the controller is CLP). This input is usually supplied

from an input DC power source. Operating voltage ranges

from GND to 28.2V.

GND (Pin 5): Ground. Internally connected to the Exposed

Pad (package paddle).

SHDN (Pin 6): Active-Low Shutdown Input. Driving SHDN

below 300mV unconditionally forces the LTC4012-3 into the

shutdown state. This input has a 40kΩ internal pull-down

to GND. Operating voltage range is GND to INTVDD.

CHRG (Pin 7): Active-Low Charge Indicator Output. This

open-drain output provides three levels of information

about charge status using a strong pull-down, 25µA weak

pull-down or high impedance. Refer to the Operation and

Applications Information sections for further details. This

output should be left floating if not used.

ICL (Pin 8): Active-Low Input Current Limit Indicator Out-

put. This open-drain output pulls to GND when the charge

current is reduced because of AC adapter input current

limiting. This output should be left floating if not used.

VFB (Pin 9): Battery Voltage Feedback Input. An external

resistor divider between FBDIV and GND with the center

tap connected to VFB programs the charger output voltage.

In constant voltage mode, this pin is nominally at 1.2085V.

Refer to the Applications Information section for complete

details on programming battery voltage. Operating voltage

range is GND to 1.25V.

FBDIV (Pin 10): Battery Voltage Feedback Resistor Divider

Source. The LTC4012-3 connects this pin to BAT when

charging is in progress. FBDIV is an open-drain PFET output

to BAT with an operating voltage range of GND to BAT.

BAT (Pin 11): Battery Pack Connection. The LTC4012-3

uses the voltage on this pin to control PWM operation when

charging. Operating voltage range is GND to CLN.

ITH (Pin 12): PWM Control Voltage and Compensation

Node. The LTC4012-3 develops a voltage on this pin to

control cycle-by-cycle peak inductor current. An external

R-C network connected to ITH provides PWM loop com-

pensation. Refer to the Applications Information section

for further details on establishing loop stability. Operating

voltage range is GND to INTVDD.

PROG (Pin 13): Charge Current Programming and Monitor-

ing Pin. An external resistance connected between PROG

and GND, along with the current sense and PWM input

resistors, programs the maximum charge current. The

voltage on this pin can also provide a linearized indicator

of charge current. Refer to the Applications Information

section for complete details on current programming and

monitoring. Operating voltage range is GND to INTVDD.

CSN (Pin 14): Charge Current Sense Negative Input.

Place an external input resistor (RIN, Figure 1) between

this pin and the negative side of the charge current sense

resistor. Operating voltage ranges from (BAT – 50mV) to

(BAT + 200mV).

CSP (Pin 15): Charge Current Sense Positive Input.

Place an external input resistor (RIN, Figure 1) between

this pin and the positive side of the charge current sense

resistor. Operating voltage ranges from (BAT – 50mV)

to (BAT + 200mV).

40123fb

LTC4012-3

pin FuncTions

BGATE (Pin 16): External Synchronous NFET Gate Control

Output. This output provides gate drive to an external NMOS

power transistor switch used for synchronous rectification

to increase efficiency in the step-down DC/DC converter.

Operating voltage is GND to INTVDD. BGATE should be

left floating if not used.

INTVDD (Pin 17): Internal 5V Regulator Output. This pin

provides a means of bypassing the internal 5V regulator

used to power the LTC4012-3 PWM FET drivers. This sup-

ply shuts down when the LTC4012-3 shuts down. Refer to

the Application Information section for details if additional

power is drawn from this pin by the application circuit.

SW (Pin 18): PWM Switch Node. The LTC4012-3 uses the

voltage on this pin as the source reference for its topside

NFET (PWM switch) driver. Refer to the Applications In-

formation section for additional PCB layout suggestions

related to this critical circuit node. Operating voltage range

is GND to CLN.

TGATE (Pin 19): External NFET Switch Gate Control Output.

This output provides gate drive to an external NMOS power

transistor switch used in the DC/DC converter. Operating

voltage range is GND to (CLN + 5V).

BOOST (Pin 20): TGATE Driver Supply Input. A bootstrap

capacitor is returned to this pin from a charge network

connected to SW and INTVDD. Refer to the Applications

Information section for complete details on circuit topol-

ogy and component values. Operating voltage ranges from

(INTVDD – 1V) to (CLN + 5V).

GND (Exposed Pad Pin 21): Ground. The package paddle

provides a single-point ground for the internal voltage

reference and other critical LTC4012-3 circuits. It should be

soldered to a suitable PCB copper ground pad for proper

electrical operation and to obtain the specified package

thermal resistance.

40123fb

LTC4012-3

block DiagraM

–

TO INTERNAL

CIRCUITS

INTERNAL

CIRCUITS

–

1.2085V

REFERENCE

REGULATOR

PWM

LOGIC

FAULT

DETECTION

C/10

DETECTION

SHUTDOWN

CONTROL

OSCILLATOR

BAT

SHUTDOWN

CHARGE

INPUT

CURRENT

LIMIT

TGATE

BOOST

ITH

PROG

CSN

CSP

GND

(PADDLE)

4012-3 BD01

BGATE

INTVDD

INTERNAL

CIRCIUTS

VFB

CHRG

ACP

FBDIV

GND

6SHDN

ICL

CLP

CLN

INFET

DCIN

–

40123fb

LTC4012-3

TesT circuiTs

–

13 12

1.2085V

TARGET

PROG VFB ITH

4012-3 TC01

LTC4012-3

0.6V

FROM ICL

(CLP = CLN)

–

LTC1055

40123fb

LTC4012-3

operaTion

Overview

The LTC4012-3 is a synchronous step-down (buck) cur-

rent mode PWM battery charger controller. The maximum

charge current is programmed by the combination of a

charge current sense resistor (RSENSE), matched input

resistors (RIN, Figure 1), and a programming resistor

(RPROG) between the PROG and GND pins. Battery voltage

is programmed with an external resistor divider between

FBDIV and GND. In addition, the PROG pin provides a

linearized voltage output of the actual charge current.

The LTC4012-3 does not have built-in charge termination

and is flexible enough for charging any type of battery

chemistry. It is a building block IC intended for use with

an external circuit, such as a microcontroller, capable of

managing the entire algorithm required for the specific

battery being charged. The LTC4012-3 features a shut-

down input and various state indicator outputs, allowing

easy and direct management by a wide range of external

(digital) charge controllers.

Shutdown

The LTC4012-3 remains in shutdown until DCIN is greater

than 5.1V and exceeds CLP by 60mV and SHDN is driven

above 1.4V. In shutdown, current drain from the battery

is reduced to the lowest possible level, thereby increasing

standby time. When in shutdown, the ITH pin is pulled to

GND and CHRG, ICL, FET gate drivers and INTVDD are all

disabled. Charging can be stopped at any time by forcing

SHDN below 300mV.

AC Present Detection

AC present is detected as soon as DCIN exceeds BAT by at

least 500mV. Charging is not enabled until this condition is

first met. After this event, charging is no longer gated by

AC present detection. If battery voltage rises due to ESR,

or DCIN droops due to current load, the PWM will remain

enabled, even with very low input overhead, unless DCIN

falls below the supply voltage on CLP.

Input PowerPath Control

The input PFET controller performs many important func-

tions. First, it monitors DCIN and enables the charger

when this input voltage is higher than the raw CLP sys-

tem supply. Next, it controls the gate of an external input

power PFET to maintain a low forward voltage drop when

charging, creating improved efficiency. It also prevents

reverse current flow through this same PFET, providing

a suitable input blocking function. Finally, it helps avoid

synchronous boost operation during invalid operating

conditions by detecting elevated CLP voltage and forcing

the charger off.

If DCIN voltage is less than CLP, then DCIN must rise

60mV higher than CLP to enable the charger and activate

the ideal diode control. The gate of the input PFET is

driven to a voltage sufficient to regulate a forward drop

between DCIN and CLP of about 25mV. If the input voltage

differential drops below this point, the FET is turned off

slowly. If the voltage between DCIN and CLP drops to less

than –25mV, the input FET is turned off in less than 6µs

to prevent significant reverse current from flowing back

through the PFET, and the charger is disabled.

Soft-Start

Exiting the shutdown state enables the charger and releases

the ITH pin. When enabled, switching will not begin until

DCIN exceeds BAT by 500mV and ITH exceeds a threshold

that assures initial current will be positive (about 5% to

25% of the maximum programmed current). To limit inrush

current, soft-start delay is created with the compensation

values used on the ITH pin. Longer soft-start times can be

realized by increasing the filter capacitor on ITH, if reduced

loop bandwidth is acceptable. The actual charge current at

the end of soft-start will depend on which loop (current,

voltage or adapter limit) is in control of the PWM. If this

current is below that required by the ITH start-up threshold,

the resulting charge current transient duration depends on

loop compensation but is typically less than 100µs.

40123fb

LTC4012-3

Bulk Charge

When soft-start is complete, the LTC4012-3 begins sourc-

ing the current programmed by the external components

connected to CSP, CSN and PROG. Some batteries may

require a small conditioning trickle current if they are heavily

discharged. As shown in the Applications Information sec-

tion, the LT4012-3 can address this need through a variety

of low current circuit techniques on the PROG pin. Once

a suitable cell voltage has been reached, charge current

can be switched to a higher, bulk charge value.

End of Charge and CHRG Output

As the battery approaches the programmed output volt-

age, charge current will begin to decrease. The open-drain

CHRG output can indicate when the current drops to 10%

of its programmed full-scale value by turning off the strong

pull-down (open-drain FET) and turning on a weak 25µA

pull-down current. This weak pull-down state is latched

until the part enters shutdown or the sensed current rises

to roughly C/6. C/10 indication will not be set if charge

current has been reduced due to adapter input current

limiting. As the charge current approaches 0A, the PWM

continues to operate in full continuous mode. This avoids

generation of audible noise, allowing bulk ceramic capaci-

tors to be used in the application.

Charge Current Monitoring

When the LTC4012-3 is charging, the voltage on the PROG

pin varies in direct proportion to the charge current. Refer-

ring to Figure 1, the nominal PROG voltage is given by:

VI R R

RµA R

PROG CHRG SENSE PROG

IN PROG

= +

• • . •11 67

Voltage tolerance on PROG is limited by the charge current

accuracy specified in the Electrical Characteristics table.

Refer to the Applications Information section on program-

ming charge current for additional details.

operaTion

Figure 1. PWM Circuit Diagram

–

11 PWM

LOGIC

OSCILLATOR

WATCHDOG

TIMER

LOOP

COMPENSATION

TGATE

SYSTEM

POWER

RIN +

–

BGATE

CSP

CSN

PROG

VFB

ITH

RPROG

FROM ICL AMP

1.2085V

RSENSE VSENSE

ICHRG

4012-3 F01

CPROG

CLOCK

LTC4012-3

CLP

BAT RD

RIN

40123fb

LTC4012-3

Adapter Input Current Limit

The LTC4012-3 can monitor and limit current from the

input DC supply, which is normally an AC adapter. When

the programmed adapter input current is reached, charge

current is reduced to maintain the desired maximum input

current. The ITH and PROG pins will reflect the reduced

charge current. This limit function avoids overloading the

DC input source, allowing the product to operate at the

same time the battery is charging without complex load

management algorithms. The battery will automatically be

charged at the maximum possible rate that the adapter will

support, given the application’s operating condition. The

LTC4012-3 can only limit input current by reducing charge

current, and in this case the charger uses nonsynchro-

nous PWM operation to prevent boosting if the average

charge current falls below about 25% of the maximum

programmed current. Note that the ICL indicator output

becomes active (low) at an adapter input current level just

slightly less than that required for the internal amplifier to

begin to assert control over the PWM loop.

Charger Status Indicator Outputs

The LTC4012-3 open-drain indicator outputs provide valu-

able information about the IC’s operating state and can

be used for a variety of purposes in applications. Table 1

summarizes the state of the indicator outputs as a function

of LTC4012-3 operation.

operaTion

Table 1. LTC4012-3 Open-Drain Indicator Outputs

CHRG ICL CHARGER STATE

Off Off No DC Input (Shutdown)

or Reverse Current

On Off Bulk Charge

25µA Off Low Current Charge or Initial

DCIN – BAT <500mV

On On Input Current Limit During Bulk

Charge

25µA On Input Current Limit During Low

Current Charge

PWM Controller

The LTC4012-3 uses a synchronous step-down architec-

ture to produce high operating efficiency. The nominal

operating frequency of 550kHz allows use of small filter

components. The following conceptual discussion of basic

PWM operation references Figure 1.

The voltage across the external charge current sense

resistor RSENSE is measured by current amplifier, CA. This

instantaneous current (VSENSE/RIN) is fed to the PROG pin

where it is averaged by an external capacitor and converted

to a voltage by the programming resistor RPROG between

PROG and GND. The PROG voltage becomes the average

charge current input signal to error amplifier, EA. EA also

receives loop control information from the battery voltage

feedback input, VFB, and the adapter input current limit

circuit.

Figure 2. PWM Waveforms

OFF

INDUCTOR

CURRENT

TOP FET

BOTTOM FET

tOFF

THRESHOLD

SET BY ITH

VOLTAGE

4012-3 F02

40123fb

LTC4012-3

The ITH output of the error amplifier is a scaled control

voltage for one input of the PWM comparator, CC. ITH

sets a peak inductor current threshold, sensed by R1, to

maintain the desired average current through RSENSE. The

current comparator output does this by switching the state

of the RS latch at the appropriate time.

At the beginning of each oscillator cycle, the PWM clock

sets the RS latch and turns on the external topside NFET

(bottom-side synchronous NFET off) to refresh the current

carried by the external inductor L1. The inductor current

and voltage across RSENSE begin to rise linearly. CA buffers

this instantaneous voltage rise and applies it to CC with

gain supplied by R1. When the voltage across R1 exceeds

the peak level set by the ITH output of EA, the top FET

turns off and the bottom FET turns on. The inductor cur-

rent then ramps down linearly until the next rising PWM

clock edge. This closes the loop and sources the correct

inductor current to maintain the desired parameter (charge

current, battery voltage, or input current). To produce a

near constant frequency, the PWM oscillator implements

the equation:

tCLP BAT

CLP kHz

OFF =–

• 550

Repetitive, closed-loop waveforms for stable PWM opera-

tion appear in Figure 2.

PWM Watchdog Timer

As input and output conditions vary, the LTC4012-3 may

need to utilize PWM duty cycles approaching 100%.

In this case, operating frequency may be reduced well

below 550kHz. An internal watchdog timer observes the

activity on the TGATE pin. If TGATE is on for more than

40µs, the watchdog activates and forces the bottom

NFET on (top NFET off) for about 100ns. This avoids a

potential source of audible noise when using ceramic

input or output capacitors and prevents the boost supply

capacitor for the top gate driver from discharging. In low

drop out operation, the actual charge current may not be

able to reach the programmed full-scale value due to the

watchdog function.

Overvoltage Protection

The LTC4012-3 also contains overvoltage detection that

prevents transient battery voltage overshoots of more than

about 6% above the programmed output voltage. When

battery overvoltage is detected, both external MOSFETs are

turned off until the overvoltage condition clears, at which

time a new soft-start sequence begins. This is useful for

properly charging battery packs that use an internal switch

to disconnect themselves for performing functions such

as calibration or pulse mode charging.

Reverse Charge Current Protection (Anti-Boost)

Because the LTC4012-3 always attempts to operate

synchronously in full continuous mode (to avoid audible

noise from ceramic capacitors), reverse average charge

current can occur during some invalid operating condi-

tions. INFET PowerPath control avoids boosting a lightly

loaded system supply during reverse operation. However,

under heavier system loads, CLP may not boost above

DCIN, even though reverse average current is flowing. In

this case a second circuit monitors indication of reverse

average current on PROG.

If either of these circuits detects boost operation, The

LTC4012-3 turns off both external MOSFETs until the

reverse current condition clears. At that point, a new

soft-start sequence begins.

operaTion

40123fb

LTC4012-3

applicaTions inForMaTion

Programming Charge Current

The formula for charge current is:

RµA

CHRG IN

SENSE PROG

=









•.– .

1 2085 11 67

The LTC4012-3 operates best with 3.01k input resistors,

although other resistors near this value can be used to

accommodate standard sense resistor values. Refer to

the subsequent discussion on inductor selection for other

considerations that come into play when selecting input

resistors RIN.

RSENSE should be chosen according to the following

equation:

RmV

SENSE MAX

=100

where IMAX is the desired maximum charge current ICHRG.

The 100mV target can be adjusted to some degree to obtain

standard RSENSE values and/or a desired RPROG value, but

target voltages lower than 100mV will cause a proportional

reduction in current regulation accuracy.

The required minimum resistance between PROG and GND

can be determined by applying the suggested expression

for RSENSE while solving the first equation given above for

charge current with ICHRG = IMAX:

RV R

V µA R

PROG MIN IN

( )

. •

. . •

1 2085

0 1 11 67

If RIN is chosen to be 3.01k with a sense voltage of 100mV,

this equation indicates a minimum value for RPROG of

26.9k. Table 6 gives some examples of recommended

charge current programming component values based

on these equations.

The resistance between PROG and GND can simply be

set with a single a resistor, if only maximum charge cur-

rent needs to be controlled during the desired charging

algorithm.

However, some batteries require a low charge current for

initial conditioning when they are heavily discharged. The

charge current can then be safely switched to a higher

level after conditioning is complete. Figure 3 illustrates

one method of doing this with 2-level control of the PROG

pin resistance. Turning Q1 off reduces the charge current

to IMAX/10 for battery conditioning. When Q1 is on, the

LTC4012-3 is programmed to allow full IMAX current for

bulk charge. This technique can be expanded through

the use of additional digital control inputs for an arbitrary

number of pre-programmed current values.

For a truly continuous range of maximum charge current

control, pulse width modulation can be used as shown

in Figure 4.

Figure 3. Programming 2-Level Charge Current

2N7002

4012-3 F03

53.6k

PROG

LTC4012-3

26.7k CPROG

4.7nF

BULK

CHARGE

PRECHARGE

Figure 4. Programming PWM Current

2N7002

4012-3 F04

PROG

LTC4012-3

RPROG RMAX

511k

CPROG

40123fb

LTC4012-3

applicaTions inForMaTion

The value of RPROG controls the maximum value of charge

current which can be programmed (Q1 continuously on).

PWM of the Q1 gate voltage changes the value of RPROG

to produce lower currents. The frequency of this modula-

tion should be higher than a few kHz, and CPROG must be

increased to reduce the ripple caused by switching Q1. In

addition, it may be necessary to increase loop compensa-

tion capacitance connected to ITH to maintain stability or

prevent large current overshoot during start-up. Selecting

a higher Q1 PWM frequency (≈10kHz) will reduce the need

to change CPROG or other compensation values. Charge

current will be proportional to the duty cycle of the PWM

input on the gate of Q1.

Programming LTC4012-3 Output Voltage

Figure 5 shows the external circuit for programming the

charger output voltage. The voltage is then governed by

the following equation:

VV R R

RR R A R B

BAT =+

( )

= +

1 2085 1 2

22 2 2

. • ,

See Table 2 for approximate resistor values for R2.

R R VR R A R B1 2 1 2085 1 2 2 2=







= +

VBAT

.– ,

Selecting R2 to be less than 50k and the sum of R1 and

R2 at least 200k or above, achieves the lowest possible

error at the VFB sense input. Note that sources of error

such as R1 and R2 tolerance, FBDIV RON or VFB input im-

pedance are not included in the specifications given in the

Electrical Characteristics. This leads to the possibility that

very accurate (0.1%) external resistors might be required.

Actually, the temperature rise of the LTC4012-3 will rarely

exceed 50°C at the end of charge, because charge current

will have tapered to a low level. This means that 0.25%

resistors will normally provide the required level of overall

accuracy. Table 2 gives recommended values for R1 and

R2 for popular lithium-ion battery voltages. For values

of R1 above 200k, addition of capacitor CZ may improve

transient response and loop stability. A value of 10pF is

normally adequate.

Table 2. Programming Output Voltage

VBAT

(V)

R1 (0.25%)

(kΩ)

R2A (0.25%)

(kΩ)

R2B (1%)*

(Ω)

4.1 165 69 –

4.2 167 67.3 200

8.2 162 28 –

8.4 169 28.4 –

12.3 301 32.8 –

12.6 294 31.2 –

16.4 284 22.6 –

16.8 271 21 –

20.5 316 19.8 –

21 298 18.2 –

24.6 298 15.4 –

25.2 397 20 –

*To Obtain Desired Accuracy Requires Series Resistors For R2.

Figure 5. Programming Output Voltage

BAT

FBDIV

85Ω

TYPICAL

VFB

LTC4012-3 R1

R2A

R2B*

4012-3 F05

GND

(EXPOSED PAD)

*OPTIONAL TRIM RESISTOR

40123fb

LTC4012-3

applicaTions inForMaTion

Programming Input Current Limit

To set the input current limit, ILIM, the minimum wall

adapter current rating must be known. To account for the

tolerance of the LTC4012-3 input current sense circuit, 5%

should be subtracted from the adapter’s minimum rated

output. Refer to Figure 6 and program the input current

limit function with the following equation:

RmV

CL LIM

=100

where ILIM is the desired maximum current draw from

the DC (adapter) input, including adjustments for tolerance,

if any.

Figure 6. Programming Input Current Limit

RCL

CDC CF 0.1µF

CLP

LTC4012-3

CLN

5.1k

4012-3 F06

10k

FROM DC

POWER

INPUT

REMAINDER

OF SYSTEM

3INFET

CLP

CLN

INTVDD

LTC4012-3 RCL

R3 = R1

2SC2412

2.49k 1%

IMX1

4012-3 F07

0.22µF

TO REMAINDER

OF SYSTEM

FROM INFET

Figure 7. Adjusting Input Current Limit

Table 3. Common RCL Values

ADAPTER

RATING

(A)

RCL VALUE (1%)

(Ω)

RCL POWER

DISSIPATION

(W)

RCL POWER

RATING

(W)

1.00 0.100 0.100 0.25

1.25 0.080 0.125 0.25

1.50 0.068 0.150 0.25

1.75 0.056 0.175 0.25

2.00 0.050 0.200 0.25

2.50 0.039 0.244 0.50

3.00 0.033 0.297 0.50

3.50 0.027 0.331 0.50

4.00 0.025 0.400 0.50

Figure 7 shows an optional circuit that can influence

the parameters of the input current limit in two ways.

The first option is to lower the power dissipation of RCL at

the expense of accuracy without changing the input current

limit value. The second is to make the input current limit

value programmable.

Often an AC adapter will include a rated current output

margin of at least +10%. This can allow the adapter cur-

rent limit value to simply be programmed to the actual

minimum rated adapter output current. Table 3 shows

some common RCL current limit programming values.

A lowpass filter formed by RF (5.1k) and CF (0.1µF) is

required to eliminate switching noise from the LTC4012-3

PWM and other system components. If input current limit-

ing is not desired, CLN should be shorted to CLP while

CLP remains connected to power.

40123fb

LTC4012-3

applicaTions inForMaTion

The overall accuracy of this circuit needs to be better

than the power source current tolerance or be margined

such that the worse-case error remains under the power

source limits.

The accuracy of the Figure 7 circuit is a function of the

INTVDD, VBE, RCL, RF

, R1 and R3 tolerances. To improve

accuracy, the tolerance of RF should be changed from

5.1k, 5% to a 2.49k 1% resistor. RCL and the programming

resistors R1 and R3 should also be 1% tolerance such

that the dominant error is INTVDD (±3%). Bias resistor R2

can be 5%. When choosing NPN transistors, both need

to have good gain (>100) at 10µA levels. Low gain NPNs

will increase programming errors. Q1 must be a matched

NPN pair. Since RF has been reduced in value by half, the

capacitor value of CF should double to 0.22µF to remain

effective at filtering out any noise.

If you wish to reduce RCL power dissipation for a given

current limit, the programming equation becomes:

mV k

CL LIM











100 5 2 49

–• .

If you wish to make the input current limit programmable,

the equation becomes:

mV k

LIM CL











100 5 2 49

–• .

The equation governing R2 for both applications is based

on the value of R1. R3 should always be equal to R1.

R2 = 0.875 • R1 Figure 8. PROG Voltage Buffer

INTVDD

PROG

<30nA

LTC4012-3

4012-3 F08

TO SYSTEM

MONITOR

–

In many notebook applications, there are situations

where two different ILIM values are needed to allow two

different power adapters or power sources to be used.

In such cases, start by setting RLIM for the high power

ILIM configuration and then use Figure 7 to set the lower

ILIM value. To toggle between the two ILIM values, take

the three ground connections shown in Figure 7, combine

them into one common connection and use a small-signal

NFET (2N7002) to open or close that common connec-

tion to circuit ground. When the NFET is off, the circuit

is defeated (floating) allowing ILIM to be the maximum

value. When the NFET is on, the circuit will become active

and ILIM will drop to the lower set value.

Monitoring Charge Current

The PROG pin voltage can be used to indicate charge cur-

rent where 1.2085V indicates full programmed current (1C)

and zero charge current is approximately equal to RPROG •

11.67µA. PROG voltage varies in direct proportion to the

charge current between this zero-current (offset) value and

1.2085V. When monitoring the PROG pin voltage, using a

buffer amplifier as shown in Figure 8 will minimize charge

current errors. The buffer amplifier may be powered from

the INTVDD pin or any supply that is always on when the

charger is on.

40123fb

LTC4012-3

applicaTions inForMaTion

Table 4. Digital Read Back State (IN, Figure 10)

LTC4012-3

CHARGER STATE

OUT STATE

Hi-Z 1

Off 1 1

C/10 Charge 0 1

Bulk Charge 0 0

Input and Output Capacitors

In addition to typical input supply bypassing (0.1µF) on

DCIN, the relatively high ESR of aluminum electrolytic ca-

pacitors is helpful for reducing ringing when hot-plugging

the charger to the AC adapter. Refer to LTC Application

Note 88 for more information.

The input capacitor between system power (drain of top

FET, Figure 1) and GND is required to absorb all input PWM

ripple current, therefore it must have adequate ripple current

rating. Maximum RMS ripple current is typically one-half

of the average battery charge current. Actual capacitance

value is not critical, but using the highest possible voltage

rating on PWM input capacitors will minimize problems.

Consult with the manufacturer before use.

Figure 10. Microprocessor Status Interface

33k

200k

4012-3 F10

VDD

3.3V

µP

OUT

LTC4012-3

CHRG 7

C/10 CHRG Indicator

The value chosen for RPROG has a strong influence on

charge current monitoring and the accuracy of the C/10

charge indicator output (CHRG). The LTC4012-3 uses the

voltage on the PROG pin to determine when charge current

has dropped to the C/10 threshold. The nominal threshold

of 400mV produces an accurate low charge current indi-

cation of C/10 as long as RPROG = 26.7k, independent of

all other current programming considerations. However,

it may sometimes be necessary to deviate from this value

to satisfy other application design goals.

If RPROG is greater than 26.7k, the actual level at which

low charge current is detected will be less than C/10. The

highest value of RPROG that can be used while reliably

indicating low charge current before reaching final VBAT

is 30.1k. RPROG can safely be set to values higher than

this, but low current indication will be lost.

If RPROG is less than 26.7k, low charge current detection

occurs at a level higher than C/10. More importantly, the

LTC4012-3 becomes increasingly sensitive to reverse cur-

rent. The lowest value of RPROG that can be used without

the risk of erroneous boost operation detection at end of

charge is 26.1k. Values of RPROG less than this should not

be used. See the Operation section for more information

about reverse current.

The nominal fractional value of IMAX at which C/10 indica-

tion occurs is given by:

mV R µA

V R

MAX

PROG

10 400 11 67

1 2085

( )

– • .

. – •• .11 67µA

( )

Direct digital monitoring of C/10 indication is possible

with an external application circuit like the one shown in

Figure 9.

By using two different value pull-up resistors, a micro-

processor can detect three states from this pin (charging,

C/10 and not charging). See Figure 10. When a digital

output port (OUT) from the microprocessor drives one

of the resistors and a second digital input port polls the

network, the charge state can be determined as shown

in Table 4.

Figure 9. Digital C/10 Indicator

INTVDD

CHRG

TP0610T

2N7002

100k

LTC4012-3

4012-3 F09

100k

VLOGIC

100k

C/10

CHRG

100k

2N7002

40123fb

LTC4012-3

The output capacitor shown across the battery and ground

must also absorb PWM output ripple current. The general

formula for this capacitor current is:

L f

RMS

BAT BAT

CLP

PWM











0 29 1

. • • –

•

For example, IRMS = 0.22A with:

VBAT = 12.6V

VCLP = 19V

L1 = 10µH

fPWM = 550kHz

High capacity ceramic capacitors (20µF or more) available

from a variety of manufacturers can be used for input/out-

put capacitors. Other alternatives include OS-CON and

POSCAP capacitors from Sanyo.

Low ESR solid tantalum capacitors have high ripple cur-

rent rating in a relatively small surface mount package,

but exercise caution when using tantalum for input or

output bulk capacitors. High input surge current can be

created when the adapter is hot-plugged to the charger

or when a battery is connected to the charger. Solid tan-

talum capacitors have a known failure mechanism when

subjected to very high surge currents. Select tantalum

capacitors that have high surge current ratings or have

been surge tested.

EMI considerations usually make it desirable to minimize

ripple current in battery leads. Adding Ferrite beads or

inductors can increase battery impedance at the nominal

550kHz switching frequency. Switching ripple current splits

between the battery and the output capacitor in inverse

relation to capacitor ESR and the battery impedance. If

the ESR of the output capacitor is 0.2Ω and the battery

impedance is raised to 4Ω with a ferrite bead, only 5%

of the current ripple will flow to the battery.

Inductor Selection

Higher switching frequency generally results in lower ef-

ficiency because of MOSFET gate charge losses, but it allows

smaller inductor and capacitor values to be used. A primary

effect of the inductor value L1 is the amplitude of ripple

current created. The inductor ripple current ∆IL decreases

with higher inductance and PWM operating frequency:

∆I

L f

BAT BAT

CLP

PWM











• –

•

Accepting larger values of ∆IL allows the use of low in-

ductance, but results in higher output voltage ripple and

greater core losses. Lower charge currents generally call

for larger inductor values.

The LTC4012-3 limits maximum instantaneous peak in-

ductor current during every PWM cycle. To avoid unstable

switch waveforms, the ripple current must satisfy:

∆ImV

SENSE

MAX

<









2150

• –

so choose:

fmV

CLP

PWM SENSE MAX

10 125

150

>









. •

• –

For C-grade parts, a reasonable starting point for setting

ripple current is ∆IL = 0.4 • IMAX. For I-grade parts, use

∆IL = 0.2 • IMAX only if the IC will actually be used to charge

batteries over the wider I-grade temperature range. The

voltage compliance of internal LTC4012-3 circuits also

imposes limits on ripple current. Select RIN (in Figure 1)

to avoid average current errors in high ripple designs. The

following equation can be used for guidance:

R I

µA RR I

µA

SENSE L IN SENSE L

• •∆ ∆

50 20

≤ ≤

applicaTions inForMaTion

40123fb

LTC4012-3

RIN should not be less than 2.37k or more than 6.04k. Val-

ues of RIN greater than 3.01k may cause some reduction in

programmed current accuracy. Use these equations and

guidelines, as represented in Table 5, to help select the cor-

rect inductor value. This table was developed for C-grade

parts to maintain maximum ∆IL near 0.6 • IMAX with fPWM at

550kHz and VBAT = 0.5 • VCLP (the point of maximum ∆IL),

assuming that inductor value could also vary by 25% at

IMAX. For I-grade parts, reduce maximum ∆IL to less than

0.4 • IMAX, but only if the IC will actually be used to charge

batteries over the wider I-grade temperature range. In that

case, a good starting point can be found by multiplying

the inductor values shown in Table 5 by a factor of 1.6 and

rounding up to the nearest standard value.

Table 5. Minimum Typical Inductor Values

VCLP

(Typ) IMAX RSENSE RIN RPROG

<10V ≥10µH 1A 100mΩ 3.01k 26.7k

10V to 20V ≥20µH 1A 100mΩ 3.01k 26.7k

>20V ≥28µH 1A 100mΩ 3.01k 26.7k

<10V ≥5.1µH 2A 50mΩ 3.01k 26.7k

10V to 20V ≥10µH 2A 50mΩ 3.01k 26.7k

>20V ≥14µH 2A 50mΩ 3.01k 26.7k

<10V ≥3.4µH 3A 33mΩ 3.01k 26.7k

10V to 20V ≥6.8µH 3A 33mΩ 3.01k 26.7k

>20V ≥9.5µH 3A 33mΩ 3.01k 26.7k

<10V ≥2.5µH 4A 25mΩ 3.01k 26.7k

10V to 20V ≥5.1µH 4A 25mΩ 3.01k 26.7k

>20V ≥7.1µH 4A 25mΩ 3.01k 26.7k

To guarantee that a chosen inductor is optimized in any

given application, use the design equations provided and

perform bench evaluation in the target application, par-

ticularly at duty cycles below 20% or above 80% where

PWM frequency can be much less than the nominal value

of 550kHz.

TGATE BOOST Supply

Use the external components shown in Figure 11 to develop

a bootstrapped BOOST supply for the TGATE FET driver.

A good set of equations governing selection of the two

capacitors is:

VC C

120

4 5 2 20 1= =

•

., •

where QG is the rated gate charge of the top external NFET

with VGS = 4.5V. The maximum average diode current is

then given by:

ID = QG • 665kHz

To improve efficiency by increasing VGS applied to the

top FET, substitute a Schottky diode with low reverse

leakage for D1.

PWM jitter has been observed in some designs operating

at higher VIN/VOUT ratios. This jitter does not substantially

affect DC charge current accuracy. A series resistor with a

value of 5Ω to 20Ω can be inserted between the cathode

of D1 and the BOOST pin to remove this jitter, if present.

A resistor case size of 0603 or larger is recommended to

lower ESL and achieve the best results.

applicaTions inForMaTion

Figure 11. TGATE Boost Supply

BOOST

INTVDD

LTC4012-3

4012-3 F11

2µF

0.1µF

L1 TO

RSENSE

1N4148

40123fb

LTC4012-3

applicaTions inForMaTion

FET Selection

Two external power MOSFETs must be selected for use

with the charger: an N-channel power switch (top FET)

and an N-channel synchronous rectifier (bottom FET).

Peak gate-to-source drive levels are internally set to

about 5V. Consequently, logic-level FETs must be used.

In addition to the fundamental DC current, selection

criteria for these MOSFETs also include channel resis-

tance RDS(ON), total gate charge QG, reverse transfer

capacitance CRSS, maximum rated drain-source voltage

BVDSS and switching characteristics such as td(ON/OFF).

Power dissipation for each external FET is given by:

PV I T R

k V

D TOP BAT MAX DS ON

CLP

( ) ( )

• •

•

( )

21δ∆

LLP MAX RSS

D BOT CLP BAT M

I C kHz

PV V I

2665• • •

– •

( ) =

( )

AAX DS ON

CLP

T R

21• ( )

( )

δ∆

where δ is the temperature dependency of RDS(ON),

∆T is the temperature rise above the point specified in

the FET data sheet for RDS(ON) and k is a constant in-

versely related to the internal LTC4012-3 top gate driver.

The term (1 + δ∆T) is generally given for a MOSFET in the

form of a normalized RDS(ON) curve versus temperature,

but δ of 0.005/°C can be used as a suitable approxima-

tion for logic-level FETs if other data is not available.

CRSS = ∆QGD/∆VDS is usually specified in the MOSFET

characteristics. The constant k = 2 can be used in esti-

mating top FET dissipation. The LTC4012-3 is designed

to work best with external FET switches with a total gate

charge at 5V of 15nC or less.

For VCLP < 20V, high charge current efficiency generally

improves with larger FETs, while for VCLP > 20V, top gate

transition losses increase rapidly to the point that using

a topside NFET with higher RDS(ON) but lower CRSS can

actually provide higher efficiency. If the charger will be

operated with a duty cycle above 85%, overall efficiency

is normally improved by using a larger top FET.

The synchronous (bottom) FET losses are greatest at high

input voltage or during a short circuit, which forces a low

side duty cycle of nearly 100%. Increasing the size of this

FET lowers its losses but increases power dissipation in the

LTC4012-3. Using asymmetrical FETs will normally achieve

cost savings while allowing optimum efficiency.

Select FETs with BVDSS that exceeds the maximum VCLP

voltage that will occur. Both FETs are subjected to this level

of stress during operation. Many logic-level MOSFETs are

limited to 30V or less.

The LTC4012-3 uses an improved adaptive TGATE and

BGATE drive that is insensitive to MOSFET inertial delays,

td(ON/OFF), to avoid overlap conduction losses. Switching

characteristics from power MOSFET data sheets apply

only to a specific test fixture, so there is no substitute for

bench evaluation of external FETs in the target application.

In general, MOSFETs with lower inertial delays will yield

higher efficiency.

Diode Selection

A Schottky diode in parallel with the bottom FET and/or

top FET in an LTC4012-3 application clamps SW during

the non-overlap times between conduction of the top and

bottom FET switches. This prevents the body diode of the

MOSFETs from forward biasing and storing charge, which

could reduce efficiency as much as 1%. One or both diodes

can be omitted if the efficiency loss can be tolerated. A 1A

Schottky is generally a good size for 3A chargers due to the

low duty cycle of the non-overlap times. Larger diodes can

actually result in additional efficiency (transition) losses

due to larger junction capacitance.

Loop Compensation and Soft-Start

The three separate PWM control loops of the LTC4012-3

can be compensated by a single set of components at-

tached between the ITH pin and GND. As shown in the

typical LTC4012-3 application, a 6.04k resistor in series

with a capacitor of at least 0.1µF provides adequate loop

compensation for the majority of applications.

40123fb

LTC4012-3

Figure 12. High Speed Switching Path

4012 F12

VBAT

L1 RSENSE

HIGH

FREQUENCY

CIRCULATING

PATH BAT

ANALOG

GROUND

SYSTEM

GROUND

SWITCH NODE

CIN

SWITCHING GROUND

COUT

VIN

GND

D1 +

The LTC4012-3 can be soft-started with the compensation

capacitor on the ITH pin. At start-up, ITH will quickly rise

to about 0.25V, then ramp up at a rate set by the com-

pensation capacitor and the 40µA ITH bias current. The

full programmed charge current will be reached when ITH

reaches approximately 2V. With a 0.1µF capacitor, the time

to reach full charge current is usually greater than 1.5ms.

This capacitor can be increased if longer start-up times

are required, but loop bandwidth and dynamic response

will be reduced.

INTVDD Regulator Output

Bypass the INTVDD regulator output to GND with a low

ESR X5R or X7R ceramic capacitor with a value of 0.47µF

or larger. The capacitor used to build the BOOST supply

(C2 in Figure 11) can serve as this bypass. Do not draw

more than 30mA from this regulator for the host system,

governed by IC power dissipation.

Calculating IC Power Dissipation

The user should ensure that the maximum rated junction

temperature is not exceeded under all operating conditions.

The thermal resistance of the LTC4012-3 package (θJA)

is 37°C/W, provided the Exposed Pad is in good thermal

contact with the PCB. The actual thermal resistance in the

application will depend on forced air cooling and other heat

sinking means, especially the amount of copper on the

PCB to which the LTC4012-3 is attached. The following

formula may be used to estimate the maximum aver-

age power dissipation, PD (in watts), of the LTC4012-3,

which is dependent upon the gate charge of the external

MOSFETs. This gate charge, which is a function of both

gate and drain voltage swings, is determined from speci-

fications or graphs in the manufacturer’s data sheet. For

the equation below, find the gate charge for each transistor

assuming 5V gate swing and a drain voltage swing equal

to the maximum VCLP voltage. Maximum LTC4012-3

power dissipation under normal operating conditions is

then given by:

PD = DCIN(3mA + IDD + 665kHz(QTGATE + QBGATE))

– 5IDD

applicaTions inForMaTion

where:

IDD = Average external INTVDD load current, if any

QTGATE = Gate charge of external top FET in Coulombs

QBGATE = Gate charge of external bottom FET in

Coulombs

PCB Layout Conciderations

To prevent magnetic and electrical field radiation and

high frequency resonant problems, proper layout of the

components connected to the LTC4012-3 is essential.

Refer to Figure 12. For maximum efficiency, the switch

node rise and fall times should be minimized. The following

PCB design priority list will help insure proper topology.

Layout the PCB using this specific order.

1. Input capacitors should be placed as close as possible

to switching FET supply and ground connections with

the shortest copper traces possible. The switching

FETs must be on the same layer of copper as the input

capacitors. Vias should not be used to make these

connections.

2. Place the LTC4012-3 close to the switching FET gate

terminals, keeping the connecting traces short to pro-

duce clean drive signals. This rule also applies to IC

supply and ground pins that connect to the switching

FET source pins. The IC can be placed on the opposite

side of the PCB from the switching FETs.

40123fb

LTC4012-3

3. Place the inductor input as close as possible to the

switching FETs. Minimize the surface area of the switch

node. Make the trace width the minimum needed to

support the programmed charge current. Use no cop-

per fills or pours. Avoid running the connection on

multiple copper layers in parallel. Minimize capacitance

from the switch node to any other trace or plane.

4. Place the charge current sense resistor immediately

adjacent to the inductor output, and orient it such that

current sense traces to the LTC4012-3 are not long.

These feedback traces need to be run together as a

single pair with the smallest spacing possible on any

given layer on which they are routed. Locate any filter

component on these traces next to the LTC4012-3,

and not at the sense resistor location.

5. Place output capacitors adjacent to the sense resistor

output and ground.

6. Output capacitor ground connections must feed into

the same copper that connects to the input capacitor

ground before connecting back to system ground.

7. Connection of switching ground to system ground,

or any internal ground plane, should be single-point.

If the system has an internal system ground plane,

a good way to do this is to cluster vias into a single

star point to make the connection.

8. Route analog ground as a trace tied back to the

LTC4012-3 GND pin and paddle before connecting

to any other ground. Avoid using the system ground

plane. A useful CAD technique is to make analog

ground a separate ground net and use a 0Ω resistor

to connect analog ground to system ground.

9. A good rule of thumb for via count in a given high

current path is to use 0.5A per via. Be consistent when

applying this rule.

10. If possible, place all the parts listed above on the same

PCB layer.

11. Copper fills or pours are good for all power connections

except as noted above in Rule 3. Copper planes on

multiple layers can also be used in parallel. This helps

with thermal management and lowers trace inductance,

which further improves EMI performance.

12. For best current programming accuracy, provide a

Kelvin connection from RSENSE to CSP and CSN. See

Figure 13 for an example.

13. It is important to minimize parasitic capacitance on

the CSP and CSN pins. The traces connecting these

pins to their respective resistors should be as short

as possible.

applicaTions inForMaTion

Figure 13. Kelvin Sensing of Charge Current

TO CSP

RIN

4012 F13

DIRECTION OF CHARGING CURRENT

RSENSE

TO CSN

RIN

40123fb

LTC4012-3

package DescripTion

4.00 ± 0.10

NOTE:

1. DRAWING IS PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220

VARIATION (WGGD-1)—TO BE APPROVED

2. DRAWING NOT TO SCALE

3. ALL DIMENSIONS ARE IN MILLIMETERS

4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE

MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE

5. EXPOSED PAD SHALL BE SOLDER PLATED

6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION

ON THE TOP AND BOTTOM OF PACKAGE

PIN 1

TOP MARK

(NOTE 6)

0.40 ± 0.10

2019

BOTTOM VIEW—EXPOSED PAD

2.00 REF

2.45 ± 0.10

0.75 ± 0.05 R = 0.115

TYP

R = 0.05

TYP

0.25 ± 0.05

0.50 BSC

0.200 REF

0.00 – 0.05

(UF20) QFN 01-07 REV A

RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS

APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED

0.70 ±0.05

0.25 ±0.05

0.50 BSC

2.00 REF 2.45 ± 0.05

3.10 ± 0.05

4.50 ± 0.05

PACKAGE OUTLINE

PIN 1 NOTCH

R = 0.20 TYP

OR 0.35 s 45°

CHAMFER

2.45 ± 0.10

2.45 ± 0.05

UF Package

20-Lead Plastic QFN (4mm × 4mm)

(Reference LTC DWG # 05-08-1710 Rev A)

40123fb

LTC4012-3

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.

However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-

tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.

revision hisTory

REV DATE DESCRIPTION PAGE NUMBER

B 3/10 I-Grade Part Added. Reflected Throughout the Data Sheet 1 to 28

(Revision history begins at Rev B)

40123fb

LTC4012-3

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com

 LINEAR TECHNOLOGY CORPORATION 2009

LT 0610 REV B • PRINTED IN USA

relaTeD parTs

PART NUMBER DESCRIPTION COMMENTS

LTC4006 Small, High Efficiency, Fixed Voltage, Lithium-Ion

Battery Chargers with Termination

Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter Current

Limit and Thermistor Sensor, 16-pin SSOP Package

LTC4007 High Efficiency, Programmable Voltage, Lithium-Ion

Battery Charger with Termination

Complete Charger for 3- or 4-Cell Li-Ion Batteries, AC Adapter Current

Limit, Thermistor Sensor and Indicator Outputs

LTC4008/LTC4008-1 High Efficiency, Programmable Voltage/Current Battery

Chargers

Constant-Current/Constant-Voltage Switching Regulator, Resistor

Voltage/Current Programming, Thermistor Sensor and Indicator

Outputs, AC Adapter Current Limit (Omitted on 4008-1)

LTC4009/LTC4009-1

LTC4009-2

High Efficiency, Multichemistry Battery Charger Constant-Current/Constant-Voltage Switching Regulator in a 20-Lead

QFN Package, AC Adapter Current Limit, Indicator Outputs

LTC4012/LTC4012-1

LTC4012-2

High Efficiency, Multi Chemistry Battery Chargers with

PowerPath Control

Constant-Current/Constant-Voltage Switching Regulator in a 20-Lead

QFN Package, AC Adaptor Current Limit PFET Input Ideal Diode Control,

3 Indicator Outputs

LTC4060 Standalone Linear NiMH/NiCd Fast Charger Complete NiMH/NiCd Charger in a Small 16-Pin Package, No Sense

Resistor or Blocking Diode Required

LTC4411 2.6A Low Loss Idea Diode No External MOSFET, Automatic Switching Between DC sources, 140mΩ

On Resistance in ThinSOT

package

LTC4412/LTC4412HV Low Loss PowerPath Controllers Very Low Loss Replacement for Power Supply ORing Diodes Using

Minimal External Complements, Operates up to 28V (36V for HV)

LTC4413 Dual 2.6A, 2.5V to 5.5V Ideal Diodes Low Loss Replacement for ORing Diodes, 100mΩ On Resistance

LTC4414 36V, Low Loss PowerPath Controller for Large PFETs Low Loss Replacement for ORing Diodes, Operates up to 36V

LTC4416 Dual Low Loss PowerPath Controllers Low Loss Replacement for ORing Diodes, Operates up to 36V, Drives

Large PFETs, Programmable, Autonomous Switching

Typical applicaTion

12.6V 4 Amp Charger

CLP

FROM

ADAPTER

15V AT 4A

BULK

CHARGE

0.1µF R8

5.1k R14

100k

R15

0Ω*

D1 7

18 D3

R12

294k

C10

10pF

25mΩ

R9 3.01k

0.1µF

2µF

4.7µH

R11

25mΩ

12.6V

Li-Ion

BATTERY

DCIN

CHRG

0.1µF

6.04k

10k

26.7k

53.6k

LTC4012-3

ICL

SHDN

ITH

PROG

CLN

BOOST

GND

INFET

TGATE

INTVDD

BGATE

TO/FROM

MCU

GND

CSP

CSN

BAT

FBDIV

VFB

10µF

POWER TO SYSTEM

TO POWER SYSTEM LOAD

WHEN ADAPTER IS NOT

PRESENT, USE

SCHOTTKY DIODE D5 OR

THE COMBINATION OF R14, R2

D6 AND Q4

18V

ZENER

0.1µF

4.7nF

10µF

R10 3.01k

R13

31.2k 40123 TA03

D3: CMDSH-3

D4: MBR230LSFT1

Q1: 2N7002

Q2, Q3: Si7218DN

Q4, Q5: Si7423DN

L1: 1HLP-2525CZER4R7M11

*: SEE TGATE BOOST SUPPLY IN

APPLICATIONS INFORMATION