LM5642MTC/NOPB - NATIONAL SEMICONDUCTOR

VIN

4.5V - 36V

SS/ON1 SS/ON2

SYNC

UV_Delay Vout1

1.3V-0.9VIN

Vout2

1.3V-0.9VIN

LM5642/LM5642X

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

LM5642/LM5642X High Voltage, Dual Synchronous Buck Converter with Oscillator

Synchronization

Check for Samples: LM5642,LM5642X

1FEATURES DESCRIPTION

The LM5642 series consists of two current mode

2• Two Synchronous Buck Regulators synchronous buck regulator controllers operating

• 180° Out of Phase Operation 180° out of phase with each other at a normal

• 200 kHz Fixed Nominal Frequency: LM5642 switching frequency of 200kHz for the LM5642 and at

375kHz for the LM5642X.

• 375 kHz Fixed Nominal Frequency: LM5642X

• Synchronizable Switching Frequency from 150 Out of phase operation reduces the input RMS ripple

current, thereby significantly reducing the required

kHz to 250 kHz for the LM5642 and 200 kHz to input capacitance. The switching frequency can be

500 kHz for the LM5642X synchronized to an external clock between 150 kHz

• 4.5V to 36V Input Range and 250 kHz for the LM5642 and between 200 kHz

• 50 µA Shutdown Current and 500 kHz for the LM5642X. The two switching

regulator outputs can also be paralleled to operate as

• Adjustable Output from 1.3V to 90% of Vin a dual-phase, single output regulator.

• 0.04% (Typical) Line and Load Regulation

Accuracy The output of each channel can be independently

adjusted from 1.3V to 90% of Vin. An internal 5V rail

• Current Mode Control with or without a Sense is also available externally for driving bootstrap

Resistor circuitry.

• Independent Enable/Soft-start Pins Allow Current-mode feedback control assures excellent line

Simple Sequential Startup Configuration. and load regulation and wide loop bandwidth for

• Configurable for Single Output Parallel excellent response to fast load transients. Current is

Operation. (See Figure 4)sensed across either the Vds of the top FET or

• Adjustable Cycle-by-cycle Current Limit across an external current-sense resistor connected

in series with the drain of the top FET.

• Input Under-voltage Lockout

• Output Over-voltage Latch Protection The LM5642 features analog soft-start circuitry that is

independent of the output load and output

• Output Under-voltage Protection with Delay capacitance making the soft-start behavior more

• Thermal Shutdown predictable and controllable than traditional soft-start

• Self Discharge of Output Capacitors when the circuits.

Regulator is OFF Over-voltage protection is available for both outputs.

• TSSOP and HTSSOP (Exposed PAD) Packages A UV-Delay pin is also available to allow delayed shut

off time for the IC during an output under-voltage

APPLICATIONS event.

• Embedded Computer Systems Typical Application Circuit

• Navigation Systems

• Telecom Systems

• Set-Top Boxes

• WebPAD

• Point Of Load Power Architectures

1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2All trademarks are the property of their respective owners.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

RSNS2

SW2

HDRV2

CBOOT2

VDD2

LDRV2

PGND

VIN

LDRV1

VDD1

CBOOT1

HDRV1

SW1

RSNS1

ON/SS2

FB2

COMP2

ILIM1

COMP1

FB1

SYNC

UVDELAY

VLIN5

SGND

KS1

ON/SS1

ILIM2

KS2

RSNS2

SW2

HDRV2

CBOOT2

VDD2

LDRV2

PGND

VIN

LDRV1

VDD1

CBOOT1

HDRV1

SW1

RSNS1

ON/SS2

FB2

COMP2

ILIM1

COMP1

FB1

SYNC

UVDELAY

VLIN5

SGND

KS1

ON/SS1

ILIM2

KS2

DAP

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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Connection Diagram

Figure 1. Top View Figure 2. Top View

PIN DESCRIPTIONS

The positive (+) Kelvin sense for the internal current sense amplifier of Channel 1. Use a separate trace to

KS1 (Pin 1) connect this pin to the current-sense point. It should be connected to VIN as close as possible to the current-

sense resistor. When no current-sense resistor is used, connect as close as possible to the drain node of the

upper MOSFET.

Current limit threshold setting for Channel 1. It sinks a constant current of 9.9 µA, which is converted to a voltage

ILIM1 (Pin 2) across a resistor connected from this pin to VIN. The voltage across the resistor is compared with either the VDS

of the top MOSFET or the voltage across the external current sense resistor to determine if an over-current

condition has occurred in Channel 1.

Compensation pin for Channel 1. This is the output of the internal transconductance error amplifier. The loop

COMP1 (Pin 3) compensation network should be connected between this pin and the signal ground, SGND (Pin 8).

Feedback input for channel 1. Connect to VOUT through a voltage divider to set the Channel 1 output voltage.

FB1 (Pin 4)

The switching frequency of the LM5642 can be synchronized to an external clock.

SYNC (Pin 5) SYNC = LOW: Free running at 200 kHz for LM5642, and at 375kHz for LM5642X. Channels are 180° out of

phase.

SYNC = HIGH: Waiting for external clock

SYNC = Falling Edge: Channel 1 HDRV pin goes high. Channel 2 HDRV pin goes high after 2.5 µs delay. The

maximum SYNC pulse width must be greater than 100 ns.

For SYNC = Low operation, connect this pin to signal ground through a 220 kΩresistor.

A capacitor from this pin to ground sets the delay time for UVP. The capacitor is charged from a 5 µA current

UV_DELAY (Pin 6) source. When UV_DELAY charges to 2.3V (typical), the system immediately latches off. Connecting this pin to

ground will disable the output under-voltage protection.

The output of an internal 5V LDO regulator derived from VIN. It supplies the internal bias for the chip and powers

VLIN5 (Pin 7) the bootstrap circuitry for gate drive. Bypass this pin to signal ground with a minimum of 4.7 µF ceramic capacitor.

The ground connection for the signal-level circuitry. It should be connected to the ground rail of the system.

SGND (Pin 8)

Channel 1 enable pin. This pin is internally pulled up to one diode drop above VLIN5. Pulling this pin below 1.2V

ON/SS1 (Pin 9) (open-collector type) turns off Channel 1. If both ON/SS1 and ON/SS2 pins are pulled below 1.2V, the whole chip

goes into shut down mode. Adding a capacitor to this pin provides a soft-start feature that minimizes inrush

current and output voltage overshoot.

Channel 2 enable pin. See the description for Pin 9, ON/SS1. May be connected to ON/SS1 for simultaneous

ON/SS2 (Pin 10) startup or for parallel operation.

Feedback input for channel 2. Connect to VOUT through a voltage divider to set the Channel 2 output voltage.

FB2 (Pin 11)

Product Folder Links: LM5642 LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

PIN DESCRIPTIONS (continued)

Compensation pin for Channel 2. This is the output of the internal transconductance error amplifier. The loop

COMP2 (Pin 12) compensation network should be connected between this pin and the signal ground SGND (Pin 8).

Current limit threshold setting for Channel 2. See ILIM1 (Pin 2).

ILIM2 (Pin 13)

The positive (+) Kelvin sense for the internal current sense amplifier of Channel 2. See KS1 (Pin 1).

KS2 (Pin 14)

The negative (-) Kelvin sense for the internal current sense amplifier of Channel 2. Connect this pin to the low side

RSNS2 (Pin 15) of the current sense resistor that is placed between VIN and the drain of the top MOSFET. When the Rds of the

top MOSFET is used for current sensing, connect this pin to the source of the top MOSFET. Always use a

separate trace to form a Kelvin connection to this pin.

Switch-node connection for Channel 2, which is connected to the source of the top MOSFET of Channel 2. It

SW2 (Pin 16) serves as the negative supply rail for the top-side gate driver, HDRV2.

Top-side gate-drive output for Channel 2. HDRV is a floating drive output that rides on the corresponding

HDRV2 (Pin 17) switching-node voltage.

Bootstrap capacitor connection. It serves as the positive supply rail for the Channel 2 top-side gate drive. Connect

CBOOT2 (Pin 18) this pin to VDD2 (Pin 19) through a diode, and connect the low side of the bootstrap capacitor to SW2 (Pin16).

The supply rail for the Channel 2 low-side gate drive. Connected to VLIN5 (Pin 7) through a 4.7Ωresistor and

VDD2 (Pin 19) bypassed to power ground with a ceramic capacitor of at least 1µF. Tie this pin to VDD1 (Pin 24).

Low-side gate-drive output for Channel 2.

LDRV2 (Pin 20)

The power ground connection for both channels. Connect to the ground rail of the system.

PGND (Pin 21)

The power input pin for the chip. Connect to the positive (+) input rail of the system. This pin must be connected

VIN (Pin 22) to the same voltage rail as the top FET drain (or the current sense resistor when used).

Low-side gate-drive output for Channel 1.

LDRV1 (Pin 23)

The supply rail for Channel 1 low-side gate drive. Tie this pin to VDD2 (Pin 19).

VDD1 (Pin 24)

Bootstrap capacitor connection. This pin serves as the positive supply rail for the Channel 1 top-side gate drive.

CBOOT1 (Pin 25) See CBOOT2 (Pin 18).

Top-side gate-drive output for Channel 1. See HDRV2 (Pin 17).

HDRV1 (Pin 26)

Switch-node connection for Channel 1. See SW2 (Pin16).

SW1 (Pin 27)

The negative (-) Kelvin sense for the internal current sense amplifier of Channel 1. See RSNS2 (Pin 15).

RSNS1 (Pin 28)

The power ground connection for both channels. Connect to the ground rail of the system. Use of multiple vias to

PGND (DAP) internal ground plane or GND layer helps to dissipate heat generated by output power.

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

ABSOLUTE MAXIMUM RATINGS(1)(2)

Voltages from the indicated pins to SGND/PGND:

VIN, ILIM1, ILIM2, KS1, KS2 −0.3V to 38V

SW1, SW2, RSNS1, RSNS2 −0.3 to (VIN + 0.3)V

FB1, FB2, VDD1, VDD2 −0.3V to 6V

SYNC, COMP1, COMP2, UV Delay −0.3V to (VLIN5 +0.3)V

ON/SS1, ON/SS2 (3) −0.3V to (VLIN5 +0.6)V

CBOOT1, CBOOT2 43V

CBOOT1 to SW1, CBOOT2 to SW2 −0.3V to 7V

LDRV1, LDRV2 −0.3V to (VDD+0.3)V

HDRV1 to SW1, HDRV2 to SW2 −0.3V

HDRV1 to CBOOT1, HDRV2 to CBOOT2 +0.3V

Power Dissipation (TA= 25°C)(4)

TSSOP 1.1W

HTSSOP 3.4W

Ambient Storage Temp. Range −65°C to +150°C

Soldering Dwell Time, Temp.(5) Wave 4 sec, 260°C

Infrared 10sec, 240°C

Vapor Phase 75sec, 219°C

ESD Rating (6) 2kV

(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for

which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test

conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance

characteristics may degrade when the device is not operated under the listed test conditions.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and

specifications.

(3) ON/SS1 and ON/SS2 are internally pulled up to one diode drop above VLIN5. Do not apply an external pull-up voltage to these pins. It

may cause damage to the IC.

(4) The maximum allowable power dissipation is calculated by using PDMAX = (TJMAX - TA)/θJA, where TJMAX is the maximum junction

temperature, TAis the ambient temperature and θJA is the junction-to-ambient thermal resistance of the specified package. The power

dissipation ratings results from using 125°C, 25°C, and 90.6°C/W for TJMAX, TA, and θJA respectively. A θJA of 90.6°C/W represents the

worst-case condition of no heat sinking of the 28-pin TSSOP. The HTSSOP package has a θJA of 29°C/W. The HTSSOP package

thermal ratings results from the IC being mounted on a 4 layer JEDEC standard board using the same temperature conditions as the

TSSOP package above. A thermal shutdown will occur if the temperature exceeds the maximum junction temperature of the device.

(5) See http://www.ti.com for other methods of soldering plastic small-outline packages.

(6) For testing purposes, ESD was applied using the human-body model, a 100pF capacitor discharged through a 1.5 kΩresistor.

OPERATING RATINGS (1)

VIN (VLIN5 tied to VIN) 4.5V to 5.5V

VIN (VIN and VLIN5 separate) 5.5V to 36V

Junction Temperature −40°C to +125°C

(1) Absolute maximum ratings indicate limits beyond which damage to the device may occur. Operating Range indicates conditions for

which the device is intended to be functional, but does not ensure specific performance limits. For ensured specifications and test

conditions, see the Electrical Characteristics. The ensured specifications apply only for the test conditions. Some performance

characteristics may degrade when the device is not operated under the listed test conditions.

Product Folder Links: LM5642 LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

ELECTRICAL CHARACTERISTICS

Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply

over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications

appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by

design, test, or statistical analysis.

Symbol Parameter Conditions Min Typ Max Units

System

ΔVOUT/VOUT Load Regulation VIN = 28V, Vcompx = 0.5V to 1.5V 0.04 %

Line Regulation 5.5V ≤VIN ≤36V, Vcompx =1.25V 0.04 %

VFB1_FB2 Feedback Voltage 5.5V ≤VIN ≤36V 1.2154 1.2364 1.2574 V

-20°C to 85°C 1.2179 1.2364 1.2549

IVIN Input Supply Current VON_SSx > 2V 1.1 2.0 mA

5.5V ≤VIN ≤36V

Shutdown (3) 50 110 µA

VON_SS1 = VON_SS2= 0V

VLIN5 VLIN5 Output Voltage IVLIN5 = 0 to 25mA, 4.70 55.30 V

5.5V ≤VIN ≤36V

VCLos Current Limit Comparator VIN = 6V ±2 ±7.0 mV

Offset (VILIMX −VRSNSX)

ICL Current Limit Sink Current 8.4 9.9 11.4 µA

Iss_SC1, Soft-Start Source Current VON_ss1 = VON_ss2 = 1.5V (on) 0.5 2.4 5.0 µA

Iss_SC2

Iss_SK1, Soft-Start Sink Current VON_ss1 = VON_ss2 = 1.5V 25.5 10 µA

Iss_SK2

VON_SS1, Soft-Start On Threshold 0.7 1.12 1.4 V

VON_SS2

VSSTO Soft-Start Timeout (4) 3.4 V

Threshold

Isc_uvdelay UV_DELAY Source Current UV-DELAY = 2V 259µA

Isk_uvdelay UV_DELAY Sink Current UV-DELAY = 0.4V 0.2 0.48 1.2 mA

VUVDelay UV_DELAY Threshold 2.3 V

Voltage

VUVP FB1, FB2, Under Voltage As a percentage of nominal output voltage 75 80.7 86 %

Protection Latch Threshold (falling edge)

Hysteresis 3.7 %

VOVP VOUT Overvoltage As a percentage measured at VFB1, VFB2 107 114 122 %

Shutdown Latch Threshold

Swx_R SW1, SW2 ON-Resistance VSW1 = VSW2 = 0.4V 420 487 560 Ω

(1) A typical is the center of characterization data measured with low duty cycle pulse tsting at TA= 25°C. Typicals are not ensured.

(2) All limits are specified. All electrical characteristics having room-temperature limits are tested during production with TA= TJ= 25°C. All

hot and cold limits are specified by correlating the electrical characteristics to process and temperature variations and applying statistical

process control.

(3) Both switching controllers are off. The linear regulator VLIN5 remains on.

(4) When SS1 and SS2 pins are charged above this voltage and either of the output voltages at Vout1 or Vout2 is still below the regulation

limit, the under voltage protection feature is initialized.

Product Folder Links: LM5642 LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

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ELECTRICAL CHARACTERISTICS (continued)

Unless otherwise specified, VIN = 28V, GND = PGND = 0V, VLIN5 = VDD1 = VDD2. Limits appearing in boldface type apply

over the specified operating junction temperature range, (-40°C to +125°C, if not otherwise specified). Specifications

appearing in plain type are measured using low duty cycle pulse testing with TA= 25°C (1),(2). Min/Max limits are specified by

design, test, or statistical analysis.

Symbol Parameter Conditions Min Typ Max Units

Gate Drive

ICBOOT CBOOTx Leakage Current VCBOOT1 = VCBOOT2 = 7V 10 nA

ISC_DRV HDRVx and LDRVx Source VCBOOT1 = VCBOOT2 = 5V, VSWx=0V, 0.5 A

Current HDRVx=LDRVx=2.5V

Isk_HDRV HDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, HDRVX 0.8 A

= 2.5V

Isk_LDRV LDRVx Sink Current VCBOOTx = VDDx = 5V, VSWx = 0V, LDRVX 1.1 A

= 2.5V

RHDRV HDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1 Ω

Resistance VSW1 = VSW2 = 0V

HDRV1 & 2 Sink On- 1.5 Ω

Resistance

RLDRV LDRV1 & 2 Source On- VCBOOT1 = VCBOOT2 = 5V, 3.1 Ω

Resistance VSW1 = VSW2 = 0V

VDD1 = VDD1 = 5V

LDRV1 & 2 Sink On- 1.1 Ω

Resistance

Oscillator and Sync Controls

5.5 ≤VIN ≤36V, LM5642 166 200 226

Fosc Oscillator Frequency kHz

5.5 ≤VIN ≤36V, LM5642X 311 375 424

Don_max Maximum On-Duty Cycle VFB1 = VFB2 = 1V, Measured at pins 96 98.9 %

HDRV1 and HDRV2

Ton_min Minimum On-Time 166 ns

SSOT_delta HDRV1 and HDRV2 Delta ON/SS1 = ON/SS2 = 2V 20 250 ns

On Time

VHS SYNC Pin Min High Input 21.52 V

VLS SYNC Pin Max Low Input 1.44 0.8 V

Error Amplifier

IFB1, IFB2 Feedback Input Bias VFB1_FIX = 1.5V, VFB2_FIX = 1.5V 80 ±200 nA

Current

Icomp1_SC, COMP Output Source VFB1_FIX = VFB2_FIX = 1V, 6127

Icomp2_SC Current VCOMP1 = VCOMP2 = 1V µA

-20°C to 85°C 18

Icomp1_SK, COMP Output Sink Current VFB1_FIX = VFB2_FIX = 1.5V and 6118

Icomp2_SK VCOMP1 = VCOMP2 = 0.5V µA

-20°C to 85°C 18

gm1, gm2 Transconductance 720 µmho

GISNS1, Current Sense Amplifier VCOMPx = 1.25V 4.2 5.2 7.5

GISNS2 (1&2) Gain

Voltage References and Linear Voltage Regulators

UVLO VLIN5 Under-voltage ON/SS1, ON/SS2 transition

Lockout from low to high 3.6 4.0 4.4 V

Threshold Rising

Product Folder Links: LM5642 LM5642X

SGND

COMP1

ON/SS1

COMP2

VDD1

VIN

VLIN5

SYNC

VDD2

UV_DELAY

IC1

LM5642

ON/SS2

C34

R28

C11

C12

R27

R23 R24

C27 C26

C20

C19

R10

R11

C4R6

D3A VDD

1 PF

100 nF

220 k:

10 nF

4.7 :

1 PF4.7 PF

8.2 nF

15 nF

8.45

13.7

Si4840DY

Si4850EY

10 PF

50V

2.8Arms

VIN

10 m:

10 nF 100 pF

12 k:

100:

100 pF

100:

100 nF

BAS40-06

4.2 PH

7 m: 330 PF

6.3V

10 m:

2.26

4.99 k:

C13

R13

R19

R20

C14

R14

R15

R16

C16

D3B VDD

C23

Si4840DY

Si4850EY

10 PF

50V

2.8Arms

VIN

10 m:

10 nF 100 pF

6.8 k:

100:

100 pF

100:

100 nF

BAS40-06

10 P+

12 m: 330 PF

6.3V

10 m:

8.25

4.99 k:

C16

C25

VDD

SYNC

Vin = 24V+10%

Vo1 = 1.8V, 7A

Vo2 = 3.3V, 4A

FB1

ILIM1

RSNS1

KS1

CBOOT1

HDRV1

SW1

LDRV1

PGND

FB2

ILIM2

RSNS2

KS2

SW2

HDRV2

CBOOT2

LDRV2

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

Figure 3. Typical 2 Channel Application Circuit

Product Folder Links: LM5642 LM5642X

IC1

LM5642

C34

R28

C11

R27

R23

C27 C26 C19

R10

R11

C4R6

D3A VDD

1 PF

100

220 k:

22 nF

4.7:

1 PF 4.7 PF

11.5 k:

Si4470DY x 2

Si4850EY

10 PF

50V

2.8Arms

VIN

10 m:

10 nF 100 pF

16.9 k:

100:

100 pF

100:

100 nF

BAS40-06

2.7 PH

4.5 m: 1000 PF

16V

22 m:

2.26

4.99 k:

C13

R13

C14

R14

R15

R16

C16

D3B VDD

C23

Si4470DY x 2

Si4850EY

10 PF

50V

2.8Arms

VIN

10 m:

10 nF 100 pF

16.9 k:

100:

100 pF

100:

100 nF

BAS40-06

2.7 PH

4.5 m: 1000 PF

16V

22 m:

C16

C25

VDD

SYNC Vo = 1.8V, 20A

27 nF

C10

1 PF

C24

1 PF

Vin = 30V

10%

SGND

COMP1

ON/SS1

COMP2

VDD1

VIN

VLIN5

SYNC

VDD2

UV_DELAY

ON/SS2

FB1

ILIM1

RSNS1

KS1

CBOOT1

HDRV1

SW1

LDRV1

PGND

ILIM2

RSNS2

KS2

SW2

HDRV2

CBOOT2

LDRV2

FB2

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

www.ti.com

Figure 4. Typical Single Channel Application Circuit

Product Folder Links: LM5642 LM5642X

Vref

ILIMx

KSx

10 PA

RSNSx

VDDx

LDRVx

PGNDx

Input Power

Supply

Shoot through

protection

sequencer

Shifter

and latch

0.50V

Corrective

ramp

SGND

OSC

200 kHz LM5642

375 kHz LM5642X

02.5 Ps

delay

To Ch2

Reset by

POR or SD

UVP

OVP

ON/SSx

ON/OFF

and

S/S

control

PWM comp

5V LDO

(Allways ON)

fault FAULT

TSD

UVLO

OVP

UVPG1

comparator

UVP

COMPx

CHx

output

error amp

Ch1 and Ch2 are identical

HDRVx

CHx

Output

S/S level Cycle

Skip

comp

SWx

UV_DELAY

ILIM

Comp

ISENSE

amp

-CBOOTx

reference

IREF Current

bias

Voltage

and

Current

generator

From another Ch.

From

another

CH.

Bias

Generator

VIN

Normal:

SS:

FBx

2 PA

VLIN5

SD Disable

Active

discharge

Rdson =

500:

PWM logic

control

SYNC

7 PA

5 PA

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

BLOCK DIAGRAM

Figure 5. Block Diagram

Product Folder Links: LM5642 LM5642X

100Ps/DIV

Io2, 2A/DIV

Vo2, 100mV/DIV

100Ps/DIV

Io1, 2A/DIV

Vo1, 100mV/DIV

20ms/DIV

UV DELAY, 1V/div

Vo1, 1V/div

Vo2, 1V/div

Io1, 5A/div

100ms/DIV

ON/SS1 and 2, 5V/div

Vo1, 1V/div

Vo2, 1V/div

4 ms/DIV

ON/SS1 and 2,

5V/div, VIN = 24V

Vo1, 2V/div

Vo2, 2V/div

Vo1, 2V/div

Vo2, 2V/div

5V/div, VIN = 36V

ON/SS1 and 2,

20ms/DIV

UV DELAY, 2V/div

ON/SS1, 2V/div

Vo1,

1V/div

Io1, 5A/div

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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TYPICAL PERFORMANCE CHARACTERISTICS

Softstart Waveforms (No-Load Both Channels) UVP Startup Waveform (VIN = 24V)

Figure 6. Figure 7.

Over-Current and UVP Shutdown (VIN = 24V, Io2 = 0A) Shutdown Waveforms (VIN = 24V, No-Load)

Figure 8. Figure 9.

Ch.1 Load Transient Response (VIN = 24V, Vo1 = 1.8V) Ch.2 Load Transient Response (VIN = 24V, Vo2 = 3.3V)

Figure 10. Figure 11.

Product Folder Links: LM5642 LM5642X

5.5 8 12 16 20 24 28 32 36

VIN (V)

VLIN5 (V)

5.095

5.09

5.085

5.08

5.075

5.07

5.065

-40 -20 0 25 50 75 100 125

TEMPERATURE (oC)

VLIN5 (V)

VIN = 5.5V

VIN = 36V

4.96

4.98

5.02

5.04

5.06

5.08

5.1

5.5 8 12 16 20 24 28 32 36

VIN

IQ (PA)

53.5

49.5

50.5

51.5

52.5

-40 -20 0 25 50 75 100

TEMPERATURE (oC)

IQ (PA)

125

100Ps/DIV

Io2, 2A/DIV

Vo2, 100mV/DIV

100Ps/DIV

Io1, 2A/DIV

Vo1, 100mV/DIV

LM5642, LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Ch. 2 Load Transient Response (VIN = 36V, Vo2 = 3.3V) Ch.1 Load Transient Response (VIN = 36V, Vo1 = 1.8V)

Figure 12. Figure 13.

Input Supply Current vs Temperature Input Supply Current vs VIN

(Shutdown Mode VIN = 28V) Shutdown Mode (25°C)

Figure 14. Figure 15.

VLIN5 vs Temperature VLIN5 vs VIN (25°C)

Figure 16. Figure 17.

Product Folder Links: LM5642 LM5642X

-40 -20 0 25 50 75 100 125

TEMPERATURE (oC)

400

450

500

550

600

650

700

750

EA gm (Pmho)

0 1 2 3 4 5 6 7

LOAD CURRENT (A)

EFFICIENCY (%)

VIN = 24V

VIN = 36V

100

-40 -20 0 25 50 75 100 125

1.232

1.2325

1.233

1.2335

1.234

1.2345

1.235

1.2355

1.236

1.2365

1.237

VREF (V)

TEMPERATURE (oC)

-40 -20 0 25 50 75 100 125

184

186

188

190

192

194

196

198

200

202

204

FREQUENCY (kHz)

TEMPERATURE (oC)

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Operating Frequency vs Temperature

FB Reference Voltage vs Temperature (VIN = 28V)

Figure 18. Figure 19.

Error Amplifier Tranconductance Gain

vs Efficiency vs Load Current Using Resistor Sense

Temperature Ch.1 = 1.8V, Ch.2 = Off

Figure 20. Figure 21.

Product Folder Links: LM5642 LM5642X

0 1 2 3 4 5

EFFICIENCY (%)

LOAD CURRENT (A)

VIN = 24V

VIN = 36V

100

0 1 2 3 4 5

EFFICIENCY (%)

LOAD CURRENT (A)

VIN = 24V

VIN = 36V

100

0 1 2 3 4 5 6 7

LOAD CURRENT (A)

EFFICIENCY (%)

VIN = 24V

VIN = 36V

100

LM5642, LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Efficiency vs Load Current Efficiency vs Load Current Using Vds Sense

Ch.2 = 3.3V, Ch.1 = Off Ch.2 = 1.8V, Ch.2 = Off

Figure 22. Figure 23.

Efficiency vs Load Current Using Vds Sense

Ch.2 = 3.3V, Ch.1 = Off

Figure 24.

Product Folder Links: LM5642 LM5642X

2PA

7PA

S>R

1.2V/

1.05V ON/OFF

comparator

-S/S level

S/S buffer

ON: 2.4PA source

Fault: 5.5PA sink

ON/SSx ONx

fault

disable

Vss = 1.5 Vo

Vin + 1

Css = Iss x tss

Vss

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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OPERATING DESCRIPTIONS

SOFT START

The ON/SS1 pin has dual functionality as both channel enable and soft start control. Referring to the soft start

block diagram is shown in Figure 25, the LM5642 will remain in shutdown mode while both soft start pins are

grounded.

In a normal application (with a soft start capacitor connected between the ON/SS1 pin and SGND) soft start

functions as follows: As the input voltage rises (note, Iss starts to flow when VIN ≥2.2V), the internal 5V LDO

starts up, and an internal 2.4 µA current charges the soft start capacitor. During soft start, the error amplifier

output voltage at the COMPx pin is clamped at 0.55V and the duty cycle is controlled only by the soft start

voltage. As the SSx pin voltage ramps up, the duty cycle increases proportional to the soft start ramp, causing

the output voltage to ramp up. The rate at which the duty cycle increases depends on the capacitance of the soft

start capacitor. The higher the capacitance, the slower the output voltage ramps up. When the corresponding

output voltage exceeds 98% (typical) of the set target voltage, the regulator switches from soft start to normal

operating mode. At this time, the 0.55V clamp at the output of the error amplifier releases and peak current

feedback control takes over. Once in peak current feedback control mode, the output voltage of the error

amplifier will travel within a 0.5V and 2V window to achieve PWM control. See Figure 26.

The amount of capacitance needed for a desired soft-start time can be approximated in the following equation:

where

• Iss = 2.4 µA for one channel and 4.8µA if the channels are paralleled

• tss is the desired soft-start time (1)

Finally,

(2)

During soft start, over-voltage protection and current limit remain in effect. The under voltage protection feature is

activated when the ON/SS pin exceeds the timeout threshold (3.4V typical). If the ON/SSx capacitor is too small,

the duty cycle may increase too rapidly, causing the device to latch off due to output voltage overshoot above the

OVP threshold. This becomes more likely in applications with low output voltage, high input voltage and light

load. A capacitance of 10 nF is recommended at each soft start pin to provide a smooth monotonic output ramp.

Figure 25. Soft-Start and ON/OFF

Product Folder Links: LM5642 LM5642X

OVP 1/2

1.13BG

in: 0.84BG

out:0.80BG

FBx

shutdown

HDRV: off

LDRV:off

UVPx

UVP 5u

AUV_DELAY

from other CH.

ONx

SS Timeout

from other CH. shutdown

latch OVP

HDRV: off

LDRV:on

OVP

latch UVP

fault

UVLO

TSD

OVPx

power on

reset

-SS:0.55V

OP:2V

0.45V

high clamp

low clamp

COMPx

LM5642, LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

Figure 26. Voltage Clamp at COMPx Pin

Figure 27. OVP and UVP

OVER VOLTAGE PROTECTION (OVP)

If the output voltage on either channel rises above 113% of nominal, over voltage protection activates. Both

channels will latch off. When the OVP latch is set, the high side FET driver, HDRVx, is immediately turned off

and the low side FET driver, LDRVx, is turned on to discharge the output capacitor through the inductor. To reset

the OVP latch, either the input voltage must be cycled, or both channels must be switched off (both ON/SS pins

pulled low).

UNDER VOLTAGE PROTECTION (UVP) AND UV DELAY

If the output voltage on either channel falls below 80% of nominal, under voltage protection activates. As shown

in Figure 27, an under-voltage event will shut off the UV_DELAY MOSFET, which will allow the UV_DELAY

capacitor to charge with 5µA (typical). If the UV_DELAY pin voltage reaches the 2.3V threshold both channels

will latch off. UV_DELAY will then be disabled and the UV_DELAY pin will return to 0V. During UVP, both the

high side and low side FET drivers will be turned off. If no capacitor is connected to the UV_DELAY pin, the UVP

latch will be activated immediately. To reset the UVP latch, either the input voltage must be cycled, or both

ON/SS pins must be pulled low. The UVP function can be disabled by connecting the UV_DELAY pin to ground.

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THERMAL SHUTDOWN

The LM5642 IC will enter thermal shutdown if the die temperature exceeds 160°C. The top and bottom FETs of

both channels will be turned off immediately. In addition, both soft start capacitors will begin to discharge through

separate 5.5 µA current sinks. The voltage on both capacitors will settle to approximately 1.1V, where it will

remain until the thermal shutdown condition has cleared. The IC will return to normal operating mode when the

die temperature has fallen to below 146°C. At this point the two soft start capacitors will begin to charge with

their normal 2.4 µA current sources. This allows a controlled return to normal operation, similar to the soft start

during turn-on. If the thermal shutdown condition clears before the voltage on the soft start capacitors has fallen

to 1.1V, the capacitors will first be discharged to 1.1V, and then immediately begin charging back up.

OUTPUT CAPACITOR DISCHARGE

Each channel has an embedded 480ΩMOSFET with the drain connected to the SWx pin. This MOSFET will

discharge the output capacitor of its channel if its channel is off, or the IC enters a fault state caused by one of

the following conditions:

1. UVP

2. UVLO

If an output over voltage event occurs, the HDRVx will be turned off and LDRVx will be turned on immediately to

discharge the output capacitors of both channels through the inductors.

BOOTSTRAP DIODE SELECTION

The bootstrap diode and capacitor form a supply that floats above the switch node voltage. VLIN5 powers this

supply, creating approximately 5V (minus the diode drop) which is used to power the high side FET drivers and

driver logic. When selecting a bootstrap diode, Schottky diodes are preferred due to their low forward voltage

drop, but care must be taken for circuits that operate at high ambient temperature. The reverse leakage of some

Schottky diodes can increase by more than 1000x at high temperature, and this leakage path can deplete the

charge on the bootstrap capacitor, starving the driver and logic. Standard PN junction diodes and fast rectifier

diodes can also be used, and these types maintain tighter control over reverse leakage current across

temperature.

SWITCHING NOISE REDUCTION

Power MOSFETs are very fast switching devices. In synchronous rectifier converters, the rapid increase of drain

current in the top FET coupled with parasitic inductance will generate unwanted Ldi/dt noise spikes at the source

node of the FET (SWx node) and also at the VIN node. The magnitude of this noise will increase as the output

current increases. This parasitic spike noise may produce excessive electromagnetic interference (EMI), and can

also cause problems in device performance. Therefore, it must be suppressed using one of the following

methods.

When using resistor based current sensing, it is strongly recommended to add R-C filters to the current sense

amplifier inputs as shown in Figure 29. This will reduce the susceptibility to switching noise, especially during

heavy load transients and short on time conditions. The filter components should be connected as close as

possible to the IC.

As shown in Figure 28, adding a resistor in series with the HDRVx pin will slow down the gate drive, thus slowing

the rise and fall time of the top FET, yielding a longer drain current transition time.

Usually a 3.3Ωto 4.7Ωresistor is sufficient to suppress the noise. Top FET switching losses will increase with

higher resistance values.

Small resistors (1-5 ohms) can also be placed in series with the CBOOTx pin to effectively reduce switch node

ringing. A CBOOT resistor will slow the rise time of the FET, whereas a resistor at HDRV will increase both rise

and fall times.

Product Folder Links: LM5642 LM5642X

Rsw

CBOOTx

4R7

HDRVx

SWx 0.1 PF

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SNVS219K –JUNE 2003–REVISED APRIL 2013

Figure 28. HDRV Series Resistor

CURRENT SENSING AND LIMITING

As shown in Figure 29, the KSx and RSNSx pins are the inputs of the current sense amplifier. Current sensing is

accomplished either by sensing the Vds of the top FET or by sensing the voltage across a current sense resistor

connected from VIN to the drain of the top FET. The advantages of sensing current across the top FET are

reduced parts count, cost and power loss.

The RDS-ON of the top FET is not as stable over temperature and voltage as a sense resistor, hence great care

must be used in layout for VDS sensing circuits. At input voltages above 30V, the maximum recommended output

current is 5A per channel.

Keeping the differential current-sense voltage below 200mV ensures linear operation of the current sense

amplifier. Therefore, the RDS-ON of the top FET or the current sense resistor must be small enough so that the

current sense voltage does not exceed 200 mV when the top FET is on. There is a leading edge blanking circuit

that forces the top FET on for at least 166ns. Beyond this minimum on time, the output of the PWM comparator

is used to turn off the top FET. Additionally, a minimum voltage of at least 50 mV across Rsns is recommended

to ensure a high SNR at the current sense amplifier.

Assuming a maximum of 200 mV across Rsns, the current sense resistor can be calculated as follows:

where

• Imax is the maximum expected load current, including overload multiplier (ie: 120%)

• Irip is the inductor ripple current (see Equation 17) (3)

The above equation gives the maximum allowable value for Rsns. Conduction losses will increase with larger

Rsns, thus lowering efficiency.

The peak current limit is set by an external resistor connected between the ILIMx pin and the KSx pin. An

internal 10 µA current sink on the ILIMx pin produces a voltage across the resistor to set the current limit

threshold which is then compared to the current sense voltage. A 10 nF capacitor across this resistor is required

to filter unwanted noise that could improperly trip the current limit comparator.

Product Folder Links: LM5642 LM5642X

10 PA

comp

ISENSE

amp

POWER

SUPPLY

100

100 pF

LIMx

KSx

RSNSx

10 nF

20m

13k

LIMx

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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Figure 29. Current Sense and Current Limit

Current limit is activated when the inductor current is high enough to cause the voltage at the RSNSx pin to be

lower than that of the ILIMx pin. This toggles the Ilim comparator, thus turning off the top FET immediately. The

comparator is disabled when the top FET is turned off and during the leading edge blanking time. The equation

for current limit resistor, Rlim, is as follows:

where

• Ilim is the load current at which the current limit comparator will be tripped (4)

When sensing current across the top FET, replace Rsns with the RDS-ON of the FET. This calculated Rlim value

specifies that the minimum current limit will not be less than Imax. It is recommended that a 1% tolerance resistor

be used.

When sensing across the top FET (VDS sensing), RDS-ON will show more variation than a current-sense resistor,

largely due to temperature variation. RDS-ON will increase proportional to temperature according to a specific

temperature coefficient. Refer to the FET manufacturer's datasheet to determine the range of RDS-ON values over

operating temperature or see the Component Selection section (Equation 27) for a calculation of maximum RDS-

ON. This will prevent RDS-ON variations from prematurely tripping the current limit comparator as the operating

temperature increases.

To ensure accurate current sensing using VDS sensing, special attention in board layout is required. The KSx and

RSNSx pins require separate traces to form a Kelvin connection at the corresponding current sense nodes. In

addition, the filter components R14, R16, C14, C15 should be removed.

INPUT UNDER VOLTAGE LOCKOUT (UVLO)

The input under-voltage lock out threshold, which is sensed via the VLIN5 internal LDO output, is 4.0V (typical).

Below this threshold, both HDRVx and LDRVx will be turned off and the internal 480ΩMOSFETs will be turned

on to discharge the output capacitors through the SWx pins. When the input voltage is below the UVLO

threshold, the ON/SS pins will sink 5mA to discharge the soft start capacitors and turn off both channels. As the

input voltage increases again above 4.0V, UVLO will be de-activated, and the device will restart through a normal

soft start phase. If the voltage at VLIN5 remains below 4.5V, but above the 4.0V UVLO threshold, the device

cannot be ensured to operate within specification.

If the input voltage is between 4.0V and 5.2V, the VLIN5 pin will not regulate, but will follow approximately 200

mV below the input voltage.

Product Folder Links: LM5642 LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

DUAL-PHASE PARALLEL OPERATION

In applications with high output current demand, the two switching channels can be configured to operate as a

two phase converter to provide a single output voltage with current sharing between the two switching channels.

This approach greatly reduces the stress and heat on the output stage components while lowering input ripple

current. The inductor ripple currents also cancel to a varying degree which results in lowered output ripple

voltage. Figure 4 shows an example of a typical two-phase circuit. Because precision current sense is the

primary design criteria to ensure accurate current sharing between the two channels, both channels must use

external sense resistors for current sensing. To minimize the error between the error amplifiers of the two

channels, tie the feedback pins FB1 and FB2 together and connect to a single voltage divider for output voltage

sensing. Also, tie the COMP1 and COMP2 together and connect to the compensation network. ON/SS1 and

ON/SS2 must be tied together to enable and disable both channels simultaneously.

EXTERNAL FREQUENCY SYNC

The LM5642 series has the ability to synchronize to external sources in order to set the switching frequency. This

allows the LM5642 to use frequencies from 150 kHz to 250 kHz and the LM5642X to use frequencies from 200

kHz to 500 kHz. Lowering the switching frequency allows a smaller minimum duty cycle, DMIN, and hence a

greater range between input and output voltage. Increasing switching frequency allows the use of smaller output

inductors and output capacitors (see Component Selection). In general, synchronizing all the switching

frequencies in multi-converter systems makes filtering of the switching noise easier.

The sync input can be from a system clock, from another switching converter in the system, or from any other

periodic signal with a logic low-level less than 1.4V and a logic high level greater than 2V. Both CMOS and TTL

level inputs are acceptable.

The LM5642 series uses a fixed delay between Channel 1 and Channel 2. The nominal switching frequency of

200kHz for the LM5642 corresponds to a switching period of 5µs. Channel 2 always turns its high-side switch on

2.5µs after Channel 1 Figure 30 (a). When the converter is synchronized to a frequency other than 200kHz, the

switching period is reduced or increased, while the fixed delay between Channel 1 and Channel 2 remains

constant. The phase difference between channels is therefore no longer 180°. At the extremes of the sync range,

the phase difference drops to 135° Figure 30 (b) and Figure 30 (c). The result of this lower phase difference is a

reduction in the maximum duty cycle of one channel that will not overlap the duty cycle of the other. As shown in

Input Capacitor Selection section, when the duty cycle D1 for Channel 1 overlaps the duty cycle D2 for Channel

2, the input rms current increases, requiring more input capacitors or input capacitors with higher ripple current

ratings. The new, reduced maximum duty cycle can be calculated by multiplying the sync frequency (in Hz) by

2.5x10-6 (the fixed delay in seconds). The same logic applies to the LM5642X. However the LM5642X has a

nominal switching frequency of 375kHz which corresponds to a period of 2.67µs. Therefore channel 2 of the

LM5642X always begins it's period after 1.33µs.

DMAX = FSYNC*2.5x10-6 (5)

At a sync frequency of 150 kHz, for example, the maximum duty cycle for Channel 1 that will not overlap

Channel 2 would be 37.5%. At 250 kHz, it is the duty cycle for Channel 2 that is reduced to a DMAX of 37.5%.

Product Folder Links: LM5642 LM5642X

FSW = 200 kHz

5 Ps

2.5 Ps

5 Ps

FSW = 150 kHz

6.67 Ps

2.5 Ps

FSW = 250 kHz

4 Ps

2.5 Ps

(a)

(b)

(c)

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

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Figure 30. Period Fixed Delay Example

Component Selection

OUTPUT VOLTAGE SETTING

The output voltage for each channel is set by the ratio of a voltage divider as shown in Figure 31. The resistor

values can be determined by the following equation:

where

• Vfb = 1.238V (6)

Although increasing the value of R1 and R2 will increase efficiency, this will also decrease accuracy. Therefore, a

maximum value is recommended for R2 in order to keep the output within .3% of Vnom. This maximum R2 value

should be calculated first with the following equation:

where

• 200nA is the maximum current drawn by FBx pin (7)

Product Folder Links: LM5642 LM5642X

48 12 16 20 24 28 32 36

VIN

VOUT

Operating Region

Vout

FBx

GND R1

LM5642, LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

Figure 31. Output Voltage Setting

Example: Vnom = 5V, Vfb = 1.2364V, Ifbmax = 200nA.

(8)

Choose 60K

(9)

The Cycle Skip and Dropout modes of the LM5642 series regulate the minimum and maximum output

voltage/duty cycle that the converter can deliver. Both modes check the voltage at the COMP pin. Minimum

output voltage is determined by the Cycle Skip Comparator. This circuitry skips the high side FET ON pulse

when the COMP pin voltage is below 0.5V at the beginning of a cycle. The converter will continue to skip every

other pulse until the duty cycle (and COMP pin voltage) rise above 0.5V, effectively halving the switching

frequency.

Maximum output voltage is determined by the Dropout circuitry, which skips the low side FET ON pulse

whenever the COMP pin voltage exceeds the ramp voltage derived from the current sense. Up to three low side

pulses may be skipped in a row before a minimum on-time pulse must be applied to the low side FET.

Figure 32 shows the range of ouput voltage (for Io = 3A) with respect to input voltage that will keep the converter

from entering either Skip Cycle or Dropout mode.

For input voltages below 5.5V, VLIN5 must be connected to Vin through a small resistor (approximately 4.7

ohm). This will ensure that VLIN5 does not fall below the UVLO threshold.

Figure 32. Output Voltage Range

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Output Capacitor Selection

In applications that exhibit large, fast load current swings, the slew rate of such a load current transient will likely

be beyond the response speed of the regulator. Therefore, to meet voltage transient requirements during worst-

case load transients, special consideration should be given to output capacitor selection. The total combined

ESR of the output capacitors must be lower than a certain value, while the total capacitance must be greater

than a certain value. Also, in applications where the specification of output voltage regulation is tight and ripple

voltage must be low, starting from the required output voltage ripple will often result in fewer design iterations.

ALLOWED TRANSIENT VOLTAGE EXCURSION

The allowed output voltage excursion during a load transient (ΔVc_s) is:

where

• ±δ% is the output voltage regulation window

• ±ε% is the output voltage initial accuracy (10)

Example: Vnom = 5V, δ% = 7%, ε% = 3.4%, Vrip = 40mV peak to peak.

(11)

MAXIMUM ESR CALCULATION

Unless the rise and fall times of a load transient are slower than the response speed of the control loop, if the

total combined ESR (Re) is too high, the load transient requirement will not be met, no matter how large the

capacitance.

The maximum allowed total combined ESR is:

(12)

Since the ripple voltage is included in the calculation of ΔVc_s, the inductor ripple current should not be included

in the worst-case load current excursion. Simply use the worst-case load current excursion for ΔIc_s.

Example: ΔVc_s = 160 mV, ΔIc_s = 3A. Then Re_max = 53.3 mΩ.

Maximum ESR criterion can be used when the associated capacitance is high enough, otherwise more

capacitors than the number determined by this criterion should be used in parallel.

MINIMUM CAPACITANCE CALCULATION

In a switch mode power supply, the minimum output capacitance is typically dictated by the load transient

requirement. If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed

value even if the maximum ESR requirement is met. The worst-case load transient is an unloading transient that

happens when the input voltage is the highest and when the current switching cycle has just finished. The

corresponding minimum capacitance is calculated as follows:

(13)

Notice it is already assumed the total ESR, Re, is no greater than Re_max, otherwise the term under the square

root will be a negative value. Also, it is assumed that L has already been selected, therefore the minimum L

value should be calculated before Cmin and after Re (see Inductor Selection below). Example: Re = 20 mΩ,

Vnom = 5V, ΔVc_s = 160 mV, ΔIc_s = 3A, L = 8 µH

(14)

Generally speaking, Cmin decreases with decreasing Re, ΔIc_s, and L, but with increasing Vnom and ΔVc_s.

Product Folder Links: LM5642 LM5642X

L = 36 - 3.3

200kHz x 1.2A x3.3

36 = 12.5PH

1.2A = 36 - 3.3

L x 200kHz x3.3

Irip = 36 - 3.3

200kHz x 5x10-6

3.3

x= 3A

Lmin = 36 - 3.3

200kHz x 36

3.3 x 0.02

.060

x= 5PH

LM5642, LM5642X

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SNVS219K –JUNE 2003–REVISED APRIL 2013

Inductor Selection

The size of the output inductor can be determined from the desired output ripple voltage, Vrip, and the

impedance of the output capacitors at the switching frequency. The equation to determine the minimum

inductance value is as follows:

(15)

In the above equation, Re is used in place of the impedance of the output capacitors. This is because in most

cases, the impedance of the output capacitors at the switching frequency is very close to Re. In the case of

ceramic capacitors, replace Re with the true impedance at the switching frequency.

Example: Vin = 36V, Vo = 3.3V, VRIP = 60 mV, Re = 20 mΩ, F = 200 kHz.

(16)

The actual selection process usually involves several iterations of all of the above steps, from ripple voltage

selection, to capacitor selection, to inductance calculations. Both the highest and the lowest input and output

voltages and load transient requirements should be considered. If an inductance value larger than Lmin is

selected, make sure that the Cmin requirement is not violated.

Priority should be given to parameters that are not flexible or more costly. For example, if there are very few

types of capacitors to choose from, it may be a good idea to adjust the inductance value so that a requirement of

3.2 capacitors can be reduced to 3 capacitors.

Since inductor ripple current is often the criterion for selecting an output inductor, it is a good idea to double-

check this value. The equation is:

(17)

Also important is the ripple content, which is defined by Irip /Inom. Generally speaking, a ripple content of less

than 50% is ok. Larger ripple content will cause too much power loss in the inductor.

Example: Vin = 36V, Vo = 3.3V, F = 200 kHz, L = 5 µH, 3A max IOUT

(18)

3A is 100% ripple which is too high.

In this case, the inductor should be reselected on the basis of ripple current.

Example: 40% ripple, 40% • 3A = 1.2A

(19)

(20)

When choosing the inductor, the saturation current should be higher than the maximum peak inductor current

and the RMS current rating should be higher than the maximum load current.

Input Capacitor Selection

The fact that the two switching channels of the LM5642 are 180° out of phase will reduce the RMS value of the

ripple current seen by the input capacitors. This will help extend input capacitor life span and result in a more

efficient system. Input capacitors must be selected that can handle both the maximum ripple RMS current at

highest ambient temperature as well as the maximum input voltage. In applications in which output voltages are

less than half of the input voltage, the corresponding duty cycles will be less than 50%. This means there will be

no overlap between the two channels' input current pulses.

Product Folder Links: LM5642 LM5642X

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The equation for calculating the maximum total input ripple RMS current for duty cycles under 50% is:

where

• I1 is maximum load current of Channel 1

• I2 is the maximum load current of Channel 2

• D1 is the duty cycle of Channel 1

• D2 is the duty cycle of Channel 2 (21)

Example: Imax_1 = 3.6A, Imax_2 = 3.6A, D1 = 0.42, and D2 = 0.275

(22)

Choose input capacitors that can handle 1.66A ripple RMS current at highest ambient temperature. In

applications where output voltages are greater than half the input voltage, the corresponding duty cycles will be

greater than 50%, and there will be overlapping input current pulses. Input ripple current will be highest under

these circumstances. The input RMS current in this case is given by:

(23)

Where, again, I1 and I2 are the maximum load currents of channel 1 and 2, and D1 and D2 are the duty cycles.

This equation should be used when both duty cycles are expected to be higher than 50%.

If the LM5642 is being used with an external clock frequency other than 200kHz, or 375 kHz for the LM5642X,

the preceding equations for input rms current can still be used. The selection of the first equation or the second

changes because overlap can now occur at duty cycles that are less than 50%. From the EXTERNAL

FREQUENCY SYNC section, the maximum duty cycle that ensures no overlap between duty cycles (and hence

input current pulses) is:

DMAX = FSYNC*2.5 x 10-6 (24)

There are now three distinct possibilities which must be considered when selecting the equation for input rms

current. The following applies for the LM5642, and also the LM5642X by replacing 200 kHz with 375 kHz:

1. Both duty cycles D1and D2are less than DMAX. In this case, the first, simple equation can always be used.

2. One duty cycle is greater than DMAX and the other duty cycle is less than DMAX. In this case, the system

designer can take advantage of the fact that the sync feature reduces DMAX for one channel, but lengthens it

for the other channel. For FSYNC < 200kHz, D1is reduced to DMAX while D2actually increases to (1-DMAX).

For FSYNC > 200kHz, D2is reduced to DMAX while D1increases to (1-DMAX). By using the channel reduced to

DMAX for the lower duty cycle, and the channel that has been increased for the higher duty cycle, the first,

simple rms input current equation can be used.

3. Both duty cycles are greater than DMAX. This case is identical to a system at 200 kHz where either duty cycle

is 50% or greater. Some overlap of duty cycles is specified, and hence the second, more complicated rms

input current equation must be used.

Input capacitors must meet the minimum requirements of voltage and ripple current capacity. The size of the

capacitor should then be selected based on hold up time requirements. Bench testing for individual applications

is still the best way to determine a reliable input capacitor value. Input capacitors should always be placed as

close as possible to the current sense resistor or the drain of the top FET. When high ESR capacitors such as

tantalum are used, a 1µF ceramic capacitor should be added as closely as possible to the high-side FET drain

and low-side FET source.

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

MOSFET Selection

BOTTOM FET SELECTION

During normal operation, the bottom FET is switching on and off at almost zero voltage. Therefore, only

conduction losses are present in the bottom FET. The most important parameter when selecting the bottom FET

is the on-resistance (RDS-ON). The lower the on-resistance, the lower the power loss. The bottom FET power loss

peaks at maximum input voltage and load current. The equation for the maximum allowed on-resistance at room

temperature for a given FET package, is:

where

• Tj_max is the maximum allowed junction temperature in the FET

• Ta_max is the maximum ambient temperature

• Rθja is the junction-to-ambient thermal resistance of the FET

• TC is the temperature coefficient of the on-resistance which is typically in the range of 4000ppm/°C (25)

If the calculated RDS-ON (MAX) is smaller than the lowest value available, multiple FETs can be used in parallel.

This effectively reduces the Imax term in the above equation, thus reducing RDS-ON. When using two FETs in

parallel, multiply the calculated RDS-ON (MAX) by 4 to obtain the RDS-ON (MAX) for each FET. In the case of three

FETs, multiply by 9.

(26)

If the selected FET has an Rds value higher than 35.3Ω, then two FETs with an RDS-ON less than 141 mΩ(4 x

35.3 mΩ) can be used in parallel. In this case, the temperature rise on each FET will not go to Tj_max because

each FET is now dissipating only half of the total power.

TOP FET SELECTION

The top FET has two types of losses: switching loss and conduction loss. The switching losses mainly consist of

crossover loss and losses related to the low-side FET body diode reverse recovery. Since it is rather difficult to

estimate the switching loss, a general starting point is to allot 60% of the top FET thermal capacity to switching

losses. The best way to precisely determine switching losses is through bench testing. The equation for

calculating the on resistance of the top FET is thus:

(27)

Example: Tj_max = 100°C, Ta_max = 60°C, Rqja = 60°C/W, Vin_min = 5.5V, Vnom = 5V, and Iload_max = 3.6A.

Product Folder Links: LM5642 LM5642X

-20dB/dec

(fp1 is at zero frequency)

-20dB/dec

FREQUENCY

GAIN (dB)

fz1 fz2

fp2

10 100 1

k10

k100

k1M

FREQUENCY

(Hz)

-60

-40

-20

GAIN

(dB)

-45

-90

-135

-180

PHASE (°)

Asymptoti

Gain

Phas

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

www.ti.com

(28)

When using FETs in parallel, the same guidelines apply to the top FET as apply to the bottom FET.

Loop Compensation

The general purpose of loop compensation is to meet static and dynamic performance requirements while

maintaining stability. Loop gain is what is usually checked to determine small-signal performance. Loop gain is

equal to the product of control-output transfer function and the feedback transfer function (the compensation

network transfer function). Generally speaking it is desirable to have a loop gain slope that is roughly -20dB

/decade from a very low frequency to well beyond the crossover frequency. The crossover frequency should not

exceed one-fifth of the switching frequency. The higher the bandwidth, the faster the load transient response

speed will be. However, if the duty cycle saturates during a load transient, further increasing the small signal

bandwidth will not help. Since the control-output transfer function usually has very limited low frequency gain, it is

a good idea to place a pole in the compensation at zero frequency, so that the low frequency gain will be

relatively large. A large DC gain means high DC regulation accuracy (i.e. DC voltage changes little with load or

line variations). The rest of the compensation scheme depends highly on the shape of the control-output plot.

Figure 33. Control-Output Transfer Function

As shown in Figure 33, the control-output transfer function consists of one pole (fp), one zero (fz), and a double

pole at fn (half the switching frequency). The following can be done to create a -20dB /decade roll-off of the loop

gain: Place the first pole at 0Hz, the first zero at fp, the second pole at fz, and the second zero at fn. The

resulting feedback transfer function is shown in Figure 34.

Figure 34. Feedback Transfer Function

Product Folder Links: LM5642 LM5642X

fP = 1

2SRO CO+1 - D - .5

2SfLCO

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

The control-output corner frequencies, and thus the desired compensation corner frequencies, can be

determined approximately by the following equations:

(29)

(30)

Since fp is determined by the output network, it will shift with loading (Ro). It is best to use a minimum Iout value

of approximately 100mA when determining the maximum Ro value.

Example: Re = 20 mΩ, Co = 100 uF, Romax = 5V/100 mA = 50Ω:

(31)

(32)

First determine the minimum frequency (fpmin) of the pole across the expected load range, then place the first

compensation zero at or below that value. Once fpmin is determined, Rc1 should be calculated using:

where

• B is the desired gain in V/V at fp (fz1)

• gm is the transconductance of the error amplifier

• R1 and R2 are the feedback resistors (33)

A gain value around 10dB (3.3v/v) is generally a good starting point.

Example: B = 3.3v/v, gm = 650m, R1 = 20 kKΩ, R2 = 60.4 kΩ:

(34)

Bandwidth will vary proportional to the value of Rc1. Next, Cc1 can be determined with the following equation:

(35)

Example: fpmin = 995 Hz, Rc1 = 20 kΩ:

(36)

The compensation network (Figure 35) will also introduce a low frequency pole which will be close to 0 Hz.

A second pole should also be placed at fz. This pole can be created with a single capacitor Cc2 and a shorted

Rc2 (see Figure 35). The minimum value for this capacitor can be calculated by:

(37)

Cc2 may not be necessary, however it does create a more stable control loop. This is especially important with

high load currents and in current sharing mode.

Example: fz = 80 kHz, Rc1 = 20 kΩ:

(38)

Product Folder Links: LM5642 LM5642X

compensation

network

CC1 CC2

RC1 RC2

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

www.ti.com

A second zero can also be added with a resistor in series with Cc2. If used, this zero should be placed at fn,

where the control to output gain rolls off at -40dB/dec. Generally, fn will be well below the 0dB level and thus will

have little effect on stability. Rc2 can be calculated with the following equation:

(39)

Figure 35. Compensation Network

PCB Layout Considerations

To produce an optimal power solution with the LM5642 series, good layout and design of the PCB are as

important as the component selection. The following are several guidelines to aid in creating a good layout.

KELVIN TRACES FOR SENSE LINES

When using the current sense resistor to sense the load current connect the KS pin using a separate trace to

VIN, as close as possible to the current-sense resistor. The RSNS pin should be connected using a separate

trace to the low-side of the current sense resistor. The traces should be run parallel to each other to give

common mode rejection. Although it can be difficult in a compact design, these traces should stay away from the

output inductor and switch node if possible, to avoid coupling stray flux fields. When a current-sense resistor is

not used the KS pin should be connected as close as possible to the drain node of the upper MOSFET and the

RSNS pin should be connected as close as possible to the source of the upper MOSFET using Kelvin traces. To

further help minimize noise pickup on the sense lines is to use RC filtering on the KS and RSNS pins.

SEPARATE PGND AND SGND

Good layout techniques include a dedicated ground plane, usually on an internal layer. Signal level components

like the compensation and feedback resistors should be connected to a section of this internal SGND plane. The

SGND section of the plane should be connected to the power ground at only one point. The best place to

connect the SGND and PGND is right at the PGND pin..

MINIMIZE THE SWITCH NODE

The plane that connects the power FETs and output inductor together radiates more EMI as it gets larger. Use

just enough copper to give low impedance to the switching currents, preferably in the form of a wide, but short,

trace run.

LOW IMPEDANCE POWER PATH

The power path includes the input capacitors, power FETs, output inductor, and output capacitors. Keep these

components on the same side of the PCB and connect them with thick traces or copper planes (shapes) on the

same layer. Vias add resistance and inductance to the power path, and have relatively high impedance

connections to the internal planes. If high switching currents must be routed through vias and/or internal planes,

use multiple vias in parallel to reduce their resistance and inductance. The power components should be kept

close together. The longer the paths that connect them, the more they act as antennas, radiating unwanted EMI.

Please see AN-1229 (literature number SNVA054) for further PCB layout considerations.

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

Table 1. Bill Of Materials for Figure 3 24V to 1.8, 3.3V LM5642

ID Part Number Type Size Parameters Qty Vendor

U1 LM5642 Dual TSSOP-28 1 TI

Synchronous

Controller

Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay

Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay

D3 BAS40-06 Schottky Diode SOT-23 40V 1 Vishay

L1 RLF12560T-4R2N100 Inductor 12.5x12.5x 6mm 4.2µH, 7mΩ10A 1 TDK

L2 RLF12545T-100M5R1 Inductor 12.5x12.5x 4.5mm 10µH, 12mΩ5.1A 1 TDK

C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK

C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 3 Vishay

C15

C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK

C6, C16 C5750X5R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK

C9, C23 6TPD330M Capacitor 7.3x4.3x 3.8mm 330µF, 6.3V, 10mΩ2 Sanyo

C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay

C13

C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay

C19 VJ1206Y822KXXAT Capacitor 1206 8.2nF 10% 1 Vishay

C20 VJ1206Y153KXXAT Capacitor 1206 15nF 10% 1 Vishay

C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK

R1 CRCW1206123J Resistor 1206 12kΩ5% 1 Vishay

R2, R6, R14, CRCW1206100J Resistor 1206 100Ω5% 1 Vishay

R16

R13 CRCW1206682J Resistor 1206 6.8kΩ12% 1 Vishay

R7, R15 WSL-2512 .010 1% Resistor 2512 10mΩ1W 2 Vishay

R8, R9, R12, CRCW1206000Z Resistor 1206 0Ω8 Vishay

R17, R18, R21,

R31, R32

R10 CRCW12062261F Resistor 1206 2.26kΩ1% 1 Vishay

R23 CRCW12068451F Resistor 1206 8.45kΩ1% 1 Vishay

R24 CRCW12061372F Resistor 1206 13.7kΩ1% 1 Vishay

R11, R20 CRCW12064991F Resistor 1206 4.99kΩ1% 2 Vishay

R19 CRCW12068251F Resistor 1206 8.25kΩ1% 1 Vishay

R27 CRCW12064R7J Resistor 1206 4.7Ω5% 1 Vishay

R28 CRCW1206224J Resistor 1206 220kΩ5% 1 Vishay

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

www.ti.com

Table 2. Bill of Materials for Figure 4 30V to 1.8V, 20A LM5642

ID Part Number Type Size Parameters Qty Vendor

U1 LM5642 Dual TSSOP-28 1 TI

Synchronou

s Controller

Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay

Q2, Q3, Q5, Q6 Si4470DY N-MOSFET SO-8 60V 4 Vishay

D3 BAS40-06 Schottky SOT-23 40V 1 Vishay

Diode

L1,L2 RLF12560T-2R7N110 Inductor 12.5x12.5x 6mm 2.7µH,4.5mΩ11.5A 2 TDK

C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK

C10, C24, C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 3 TDK

C6, C16, C28, C5750X5R1H106M Capacitor 2220 10µF 50V, 2.8A 4 TDK

C30

C9, C23 16MV1000WX Capacitor 10mm D20mm H 1000µF, 16V, 22mΩ2 Sanyo

C2, C13 VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 2 Vishay

C11 VJ1206Y223KXXAT Capacitor 1206 22nF, 25V 1 Vishay

C7,C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay

C19 VJ1206Y273KXXAT Capacitor 1206 27nF 10% 1 Vishay

C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK

R1, R13 CRCW1206123J Resistor 1206 16.9kΩ1% 1 Vishay

R2, R6, R14, CRCW1206100J Resistor 1206 100Ω5% 1 Vishay

R16

R7, R15 WSL-2512 .010 1% Resistor 2512 10mΩ1W 2 Vishay

R8, R9, R12, CRCW1206000Z Resistor 1206 0Ω8 Vishay

R17, R18, R21,

R31, R32

R10 CRCW12062261F Resistor 1206 2.26kΩ1% 1 Vishay

R11 CRCW12064991F Resistor 1206 4.99kΩ1% 1 Vishay

R23 CRCW12061152F Resistor 1206 11.5kΩ1% 1 Vishay

R27 CRCW12064R7J Resistor 1206 4.7Ω5% 1 Vishay

R28 CRCW1206224J Resistor 1206 220kΩ5% 1 Vishay

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

Table 3. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=1.5V,1.8V, 5A LM5642X

ID Part Number Type Size Parameters Qty Vendor

U1 LM5642X Dual TSSOP-28 1 TI

Synchronous

Controller

Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay

Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay

D3 BA54A Schottky Diode SOT-23 30V 1 Vishay

L1, L2 RLF12545T-4R2N100 Inductor 12.5x12.5x 4.5mm 4.2µH, 7mΩ6.5A 2 TDK

C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK

C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 4 Vishay

C15

C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK

C6, C28 C5750X7R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK

C9, C23 C4532X7R0J107M Capacitor 1812 100µF, 6.3V, 1mΩ2 TDK

C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay

C13

C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay

C18, C20 VJ1206Y473KXXAT Capacitor 1206 47nF 10% 2 Vishay

C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK

R1, R13 CRCW12061912F Resistor 1206 19.1kΩ1% 2 Vishay

R2, R6, R14, CRCW1206100J Resistor 1206 100Ω5% 1 Vishay

R16

R7, R15 WSL-1206 .020 1% Resistor 1206 20mΩ1W 2 Vishay

R8, R9, R12, CRCW1206000Z Resistor 1206 0Ω8 Vishay

R17, R18, R21,

R31, R32

R10, R19 CRCW12061001F Resistor 1206 1kΩ1% 2 Vishay

R11 CRCW12062611F Resistor 1206 2.61kΩ1% 1 Vishay

R20 CRCW12062321F Resistor 1206 2.32kΩ1% 1 Vishay

R22, R24 CRCW12063011F Resistor 1206 3.01kΩ1% 2 Vishay

R27 CRCW12064R7J Resistor 1206 4.7Ω5% 1 Vishay

R28 CRCW1206224J Resistor 1206 220kΩ5% 1 Vishay

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

SNVS219K –JUNE 2003–REVISED APRIL 2013

www.ti.com

Table 4. Bill Of Materials Based on Figure 3 Vin= 9-16V, VO1,2=3.3V,5V, 5A LM5642X

ID Part Number Type Size Parameters Qty Vendor

U1 LM5642X Dual TSSOP-28 1 TI

Synchronous

Controller

Q1, Q4 Si4850EY N-MOSFET SO-8 60V 2 Vishay

Q2, Q5 Si4840DY N-MOSFET SO-8 40V 2 Vishay

D3 BA54A Schottky Diode SOT-23 30V 1 Vishay

L1, L2 RLF12545T-5R6N6R1 Inductor 12.5x12.5x 4.5mm 5.6µH, 9mΩ6.1A 2 TDK

C1 C3216X7R1H105K Capacitor 1206 1µF, 50V 1 TDK

C3, C4, C14, VJ1206Y101KXXAT Capacitor 1206 100pF, 25V 4 Vishay

C15

C27 C2012X5R1C105K Capacitor 0805 1µF, 16V 1 TDK

C6, C28 C5750X7R1H106M Capacitor 2220 10µF 50V, 2.8A 2 TDK

C9, C23 C4532X7R0J107M Capacitor 1812 100µF, 6.3V, 1mΩ2 TDK

C2, C11, C12, VJ1206Y103KXXAT Capacitor 1206 10nF, 25V 4 Vishay

C13

C7, C25, C34 VJ1206Y104KXXAT Capacitor 1206 100nF, 25V 3 Vishay

C18, C20 VJ1206Y393KXXAT Capacitor 1206 39nF 10% 2 Vishay

C26 C3216X7R1C475K Capacitor 1206 4.7µF 25V 1 TDK

R1, R13 CRCW12061912F Resistor 1206 19.1kΩ1% 2 Vishay

R2, R6, R14, CRCW1206100J Resistor 1206 100Ω5% 1 Vishay

R16

R7, R15 WSL-1206 .020 1% Resistor 1206 20mΩ1W 2 Vishay

R8, R9, R12, CRCW1206000Z Resistor 1206 0Ω8 Vishay

R17, R18, R21,

R31, R32

R10, R19 CRCW12061002F Resistor 1206 10kΩ1% 2 Vishay

R11 CRCW12066191F Resistor 1206 6.19kΩ1% 1 Vishay

R20 CRCW12063321F Resistor 1206 3.32kΩ1% 1 Vishay

R22, R24 CRCW12063831F Resistor 1206 3.83kΩ1% 2 Vishay

R27 CRCW12064R7J Resistor 1206 4.7Ω5% 1 Vishay

R28 CRCW1206224J Resistor 1206 220kΩ5% 1 Vishay

Product Folder Links: LM5642 LM5642X

LM5642, LM5642X

www.ti.com

SNVS219K –JUNE 2003–REVISED APRIL 2013

REVISION HISTORY

Changes from Revision J (April 2013) to Revision K Page

• Changed layout of National Data Sheet to TI format .......................................................................................................... 32

Product Folder Links: LM5642 LM5642X

PACKAGE OPTION ADDENDUM

www.ti.com 11-Jan-2021

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead finish/

Ball material

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM5642MH/NOPB ACTIVE HTSSOP PWP 28 48 RoHS & Green SN Level-1-260C-UNLIM LM5642

LM5642MHX/NOPB ACTIVE HTSSOP PWP 28 2500 RoHS & Green SN Level-1-260C-UNLIM LM5642

LM5642MTC NRND TSSOP PW 28 48 Non-RoHS

& Green Call TI Call TI -40 to 125 LM5642

MTC

LM5642MTC/NOPB ACTIVE TSSOP PW 28 48 RoHS & Green SN Level-3-260C-168 HR -40 to 125 LM5642

MTC

LM5642MTCX NRND TSSOP PW 28 2500 Non-RoHS

& Green Call TI Call TI -40 to 125 LM5642

MTC

LM5642MTCX/NOPB ACTIVE TSSOP PW 28 2500 RoHS & Green SN Level-3-260C-168 HR -40 to 125 LM5642

MTC

LM5642XMH/NOPB ACTIVE HTSSOP PWP 28 48 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5642

XMH

LM5642XMHX/NOPB ACTIVE HTSSOP PWP 28 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5642

XMH

LM5642XMT/NOPB ACTIVE TSSOP PW 28 48 RoHS & Green SN Level-3-260C-168 HR -40 to 125 LM5642

XMT

LM5642XMTX/NOPB ACTIVE TSSOP PW 28 2500 RoHS & Green SN Level-3-260C-168 HR -40 to 125 LM5642

XMT

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may

reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.

Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

PACKAGE OPTION ADDENDUM

www.ti.com 11-Jan-2021

Addendum-Page 2

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM5642MHX/NOPB HTSSOP PWP 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1

LM5642MTCX TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1

LM5642MTCX/NOPB TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1

LM5642XMHX/NOPB HTSSOP PWP 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1

LM5642XMTX/NOPB TSSOP PW 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 29-Sep-2019

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM5642MHX/NOPB HTSSOP PWP 28 2500 367.0 367.0 35.0

LM5642MTCX TSSOP PW 28 2500 367.0 367.0 38.0

LM5642MTCX/NOPB TSSOP PW 28 2500 367.0 367.0 38.0

LM5642XMHX/NOPB HTSSOP PWP 28 2500 367.0 367.0 35.0

LM5642XMTX/NOPB TSSOP PW 28 2500 367.0 367.0 38.0

PACKAGE MATERIALS INFORMATION

www.ti.com 29-Sep-2019

Pack Materials-Page 2

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PACKAGE OUTLINE

TYP

6.6

6.2

1.1 MAX

26X 0.65

28X 0.30

0.19

8.45

TYP

0.20

0.09

0 - 8

0.10

0.02

5.65

5.25

3.15

2.75

(1)

0.25

GAGE PLANE

0.7

0.5

NOTE 3

9.8

9.6

NOTE 4

4.5

4.3

4214870/A 10/2014

PowerPAD - 1.1 mm max heightPWP0028A

PLASTIC SMALL OUTLINE

NOTES:

1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing

per ASME Y14.5M.

2. This drawing is subject to change without notice.

3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not

exceed 0.15 mm, per side.

4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm, per side.

5. Reference JEDEC registration MO-153, variation AET.

PowerPAD is a trademark of Texas Instruments.

128

0.1 C A B

PIN 1 ID

AREA

SEATING PLANE

0.1 C

SEE DETAIL A

DETAIL A

TYPICAL

SCALE 1.800

THERMAL

PAD

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EXAMPLE BOARD LAYOUT

0.05 MAX

ALL AROUND

0.05 MIN

ALL AROUND

28X (1.3)

(6.1)

28X (0.45)

(0.9) TYP

28X (1.5)

28X (0.45)

26X

(0.65)

(3)

(3.4)

NOTE 9

(5.5)

SOLDER

MASK

OPENING

(9.7)

(1.3)

(1.3) TYP

(5.8)

( ) TYP

VIA

0.2

(0.65) TYP

4214870/A 10/2014

PowerPAD - 1.1 mm max heightPWP0028A

PLASTIC SMALL OUTLINE

SOLDER MASK

DEFINED PAD

LAND PATTERN EXAMPLE

SCALE:6X

HV / ISOLATION OPTION

0.9 CLEARANCE CREEPAGE

OTHER DIMENSIONS IDENTICAL TO IPC-7351

NOTES: (continued)

6. Publication IPC-7351 may have alternate designs.

7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.

8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature

numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).

9. Size of metal pad may vary due to creepage requirement.

METAL

SOLDER MASK

OPENING

NON SOLDER MASK

DEFINED

SOLDER MASK DETAILS

SOLDER MASK

OPENING

METAL UNDER

SOLDER MASK

DEFINED

SYMM

SEE DETAILS

14 15

METAL COVERED

BY SOLDER MASK

SOLDER

MASK

OPENING

IPC-7351 NOMINAL

0.65 CLEARANCE CREEPAGE

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EXAMPLE STENCIL DESIGN

28X (1.3)

28X (0.45)

(6.1)

28X (1.5)

28X (0.45)

26X (0.65)

(3)

(5.5)

BASED ON

0.127 THICK

STENCIL

(5.8)

4214870/A 10/2014

PowerPAD - 1.1 mm max heightPWP0028A

PLASTIC SMALL OUTLINE

2.66 X 4.770.178

2.88 X 5.160.152

3.0 X 5.5 (SHOWN)0.127

3.55 X 6.370.1

SOLDER STENCIL

OPENING

STENCIL

THICKNESS

SEE TABLE FOR

DIFFERENT OPENINGS

FOR OTHER STENCIL

THICKNESSES

NOTES: (continued)

10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate

design recommendations.

11. Board assembly site may have different recommendations for stencil design.

SOLDER PASTE EXAMPLE

EXPOSED PAD

100% PRINTED SOLDER COVERAGE AREA

SCALE:6X

HV / ISOLATION OPTION

0.9 CLEARANCE CREEPAGE

OTHER DIMENSIONS IDENTICAL TO IPC-7351

SYMM

14 15

BASED ON

0.127 THICK

STENCIL BY SOLDER MASK

METAL COVERED

IPC-7351 NOMINAL

0.65 CLEARANCE CREEPAGE

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